up-to-date electronics for lab and leisure
ELBHTOr ST
108
87-
April 1975 35p
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selektor
Under this heading elektor mill be publishing selected items about nem technologies and developments.
tap preamp (part 1)
Elektor readers will by now be familiar with the TAP or Touch Activated Programmer. For reliability and ease
of operation all the preamplifier functions are controlled by TAP's and mechanical switches and potentiometers
are eliminated.
pll systems
It is the intention of this article to give an introduction to Phase- Locked- Loop (PLL) systems, without assuming
any advanced mathematical knowledge on behalf of the reader, nor any familiarity with the subject.
decimal to bed converter — J. Wittje
fido
Fido is a new electronic game in which an unfortunate dog is called by four masters at the same time. If one of
the players succeeds in getting Fido into his kennel, the game is decided: Fido stays where he is.
elektor services to readers
time machine - H.U. Heinz
Slowly-developing technological processes or natural events cannot be perceived because the eye is generally not
able to distinguish the separate stages. Such events and processes can, however, be visualised by means of
cinematographic time compression.
marine diesel
noise generator — J. Jacobs
minidrum (part 2)
The Minidrum described in the previous issue may, by the addition of various extra circuits, be extended to a
comprehensive manual drum kit. Some new instruments, a three channel ruffle system and an automatic bass-
drum are described in this article.
compressor
Compressors are now being used on an ever-increasing scale. They may be found in tape recorders, intercom
systems and baby alarms, public address systems, discotheques and of course broadcast transmitters. A com-
pressor supplements a manual volume control and allows a system to adjust itself to a wide range of input
signals with little distortion.
The design described here should find a wide range of applications with the electronics enthusiast.
disc preamp
A preamplifier-equaliser for magnetic pickup cartridges has to meet quite exacting requirements. The well-known
two-transistor configuration, operating from a 12 ... 18 V supply, invariably falls short on gain and overdrive-
margin - unless it is designed for a low nominal output voltage (about 30 mV). An alternative approach is to
make use of a good integrated amplifier. The design described here, which meets all the requirements,
employs a SN 76131 . An almost identical I.C. is the ftA 739.
a/d converter
The necessity to convert a voltage to a frequency such that the frequency is accurately proportional to the
voltage is one which arises in many different electronic systems. Some digital voltmeters use this principle.
The circuit described here is relatively simple, but nonetheless the absolute error is less than 1 .5 mV and the
relative error less than 1 % over an input voltage range of 7 mV to 2.5 V.
led displays
In response to numerous requests from readers we publish this LED-display chart to enable constructors to
find their way through the jungle of seven-segment display data.
modulation systems (part 2)
Th
410 — elektor april 1975
tap-preamp
parti
tap
preamp
Figure 1. Touch panels for the TAP's. The
contact surfaces and legends are nickel plated
with a black background.
Figure 2. The circuit of the four-position TAP.
Touching one of the input contacts causes the
corresponding output to become 'V and all
other outputs to become 'O'.
Figure 3. Circuit to show the principle of an
electronic 'make' contact. The LED indicates
that the contact is 'closed'.
Figure 4. Extension of the circuit of figure 3
to control two channels.
In a previous issue (elektor no. 2), in the article entitled 'Sonant', a new
design of audio preamplifier and control unit was discussed, which would
complement the power amplifier/loudspeaker combination of the
Sonant. This article describes the design and construction of such a
'Pre-sonant', which combines high performance with simplicity of
operation.
Figure 5. The make contact applied to a four-
preset-level volume control. The values of
R 15‘ R 22 determine the four preset volume
Figure 6. The ele ctr onic 'break' contact. When
a ‘1 * appears at input Q„ T^ and T 2 are cut off
and the LED lights to show that the contact
is 'open'.
Elektor readers will by now be familiar
with the TAP or Touch Activated Pro-
grammer. For reliability and ease of
operation all the preamplifier functions
are controlled by TAP’s and mechanical
switches and potentiometers are elimin-
ated. This necessarily leads to some
simplification of control functions, as
such things as volume and tone control
can now be implemented only in discrete
steps. This is perhaps no bad thing, as
the front panels of some modem ampli-
fiers look like something from ‘Star Trek’
and one wonders if a training course is
necessary to operate them. This design
is, therefore, not suitable for the dedi-
cated knob twiddler!
Assuming that the recording engineer has
done his job properly, many control
functions may be removed from the
front panel of the preamp and may be
replaced by internal presets. This applies
to balance and tone controls, which may
be adjusted to suit room acoustics and
personal taste, after which no further
adjustment should be necessary. The
number of control functions was thus
reduced to the following:
Input Selection: Disc, Radio, Tape,
Auxiliary.
Four preset levels.
Four settings from
mono to ‘extreme
stereo’.
Bass lift, ‘Presence’,
Rat, Treble cut.
It is hoped in a later article to include a
touch station selector for radio. The
layout of the touch panels is shown in
figure 1. These are available from the
Elektor Print Service.
Four Position TAP
All the controls mentioned above are
based on the four-position TAP shown
in figure 2, which is designed around an
RCA COSMOS IC type CD4011AE, a
quad two-input NAND gate. The circuit
operates as follows:
When the circuit is first switched on the
output of one of the gates will set to ‘1’
and all the others are held at ‘0’ since a
‘1’ is applied to their inputs via the input
Volume:
Image Width:
Tone:
resistors connected to +Vj, and via the
diodes from the output of the gate whose
output is M’. Which output sets to ‘1’ on
initial switch on is determined by the
switching speed of the individual gates
and the various resistor tolerances.
Suppose now that input 1 is touched.
Pin 1 of gate Ni is now held at ‘0’ by the
skin resistance, the output therefore be-
comes ‘1’. This ‘1’ is applied to the
inputs of the other three gates via D4 ,
D 7 and Dio respectively. Since the other
input of each of these gates is already
at ‘1’ via the input resistors R 4 , R s , R 7 ,
R g , R 10 and R n the output of N 2 -N 4
becomes ‘O’. The logic level on the anodes
of D! , D 2 and D 3 becomes ‘0’ and pin 2
of N! is held at ‘0’ by R 3 . Thus when
input 1 is released the output of N ! re-
mains at ‘ 1 ’. This explanation applies for
all the other inputs. Only one output
can be a ‘1’ at any time.
The TAP is used to control two types of
electronic switch, a make contact, as
shown in figure 3 and a break contact as
shown in figure 6. When a ‘1’ is applied
to the Q x input in figure 3, Tj is turned
on. Current flows through the LED and
resistor into the base of T 2 , which is also
turned on. The LED lights to indicate
that this switch position is activated.
The modifications necessary to switch
two channels are shown in figure 4. Ti is
now used to switch two transistors and
the base resistors are doubled in value
(within the limits of preferred resistor
values) to keep the LED current the
The Break Contact
The circuit of figure 6 operates in an
inverse manner to that of figure 4. When
the Q x input is at ‘0’ Tj is turned off.
However, T 2 and T 3 are turned on by
current flowing into their bases via the
LED, R 2 , R 3 and 114. The ‘contact’ is
thus normally ‘closed’. When a ‘1’ is
applied to the Q x input T 1 is turned on
thus grounding the bases of T 2 and T 3
and turning them off. Current flows
through the LED via R 2 and Tj so that
it lights.
As an example of the use of the make
contact a four-setting volume control is
shown in figure 5 . For the left channel
R13 and R 15 -R 2 i comprise a potentio-
meter, likewise R 14 and Ri 6 -Rm for the
right channel. When one of the inputs
Q1-Q4 is high then the corresponding
transistorsT s /T 6 -Tii/T| 2 are turned on,
grounding one end of the corresponding
collector resistor R 15 /R 16 -R 21 /R 22 • The
attenuation depends on the value of the
resistor that is grounded and may be
varied to suit personal taste. After attenu-
ation the signal is fed into the base of
T,3 (T 14 ) and the output is taken from
the collector. This and the other control
circuits will be discussed in greater detail
in next month’s article. H
pll systems
behalf of the reader, nor any
familiarity with the subject.
The need for such an introduction stems from the ever-increasing use of PLL circuits in consumer electronics
and from the increasing complexity of these circuits, which is threatening to make new developments in this
field incomprehensible to many electronics enthusiasts. The article also deals with Feedback PLL systems,
which are in many ways superior to conventional PLL circuits.
A simple receiver using the Feedback PLL principle will be described in a future issue.
It is the intention of this article
to give an introduction to Phase-
Locked-Loop (PLL) systems,
without assuming any advanced
mathematical knowledge on
A phase-locked-loop is a control system
in which an electrical quantity is con-
trolled by the phase difference between
two signals. Figure 1 shows a block dia-
gram of an arbitrary servo control sys-
tem.
Ax and Ay are quantities of the same
form such as A.C. or D.C. potentials.
These quantities are compared with one
another in block C by, for example,
multiplication or subtraction. The result
of the comparison is processed in block C
in such a way that quantity Ay is ad-
justed. The form of processing deter-
mines a number of the control charac-
teristics such as the control time
constant. Quantity Ay is readjusted in
such a way that a state of equilibrium is
reached at the output of C.
Figure 2 is a block diagram of a PLL. In
this case control is based on the phase
difference between the input signal (1)
and the signal (2) from a Voltage-
Controlled Oscillator (VCO) so the con-
tents of block <j> must be able to recognize
this difference.
The VCO is controlled in such a way
that a specific phase difference is main-
tained between the output from the VCO
and the input signal. The speed with
which the PLL adjusts the VCO to follow
any change in the input signal depends,
in the first instance, on the characteristics
of the low-pass filter LPF.
When two signals are multiplied together,
the product includes a component that
is proportional to their phase difference
and that can be filtered out from the
other components. Block <t> performs this
multiplication. In practical circuits the
input signal is multiplied by a square-
wave output from the VCO, which means
in effect that alternate half cycles of the
VCO square wave multiply the input
signal by +1 and —1. The waveforms in
figure 3 should make it easier to under-
stand the mode of operation.
In figure 3a the input (represented as a
sinusoid) is shown and below it a VCO
square wave of the same frequency is
repeated with phase relationships varying
progressively from in-phase to 1 80 lead-
ing (figures 3b, 3d, 3f, 3h and 3j). During
the positive half-cycles of the VCO
square wave (in any particular phase) the
associated ‘product’ waveform (fig-
ures 3c, 3e, 3g, 3i and 3k) is the same as
the input sine wave of 3a. During the
negative half-cycles of the square wave
the sine wave of 3a is polarity-changed
in the product waveform. This is equiv-
alent to multiplying the two waveforms
together.
In the first product waveform (3c), which
is associated with the in-phase square
wave 3b, it will be seen that the product
never becomes negative, in fact it is a
full-wave rectified version of the sine
wave. Its filtered D.C. value is thus
unmistakably positive. When the square
wave is leading by 45°, as in 3d, the
product 3e clearly has a greater area
above the line than below. Its mean
D.C. level is therefore also positive, but
less than 3c. When the square wave leads
by 90°, as in 3f, the product 3g has equal
areas above and below the line, so its
D.C. value is zero. With leads greater
than 90° the D.C. value of the product
becomes negative, reaching a maximum
(negative) value at +180° (3h to 3k).
Summarising; the D.C. value of the prod-
uct waveform varies from a maximum
positive value when the square wave is
in phase with the input signal, through
zero when the square wave leads by 90°,
to a maximum negative value when the
square wave leads by 180°.
Assume now that the input and VCO
frequencies are precisely equal and that
the PLL is locked in (ignoring, for the
moment, how it got that way). The
VCO square wave will be leading th
input signal by 90° and the D.C. outpi
of the phase comparator (multiplier) w
be zero. Suppose now that the VCO fn
quency tends to increase. The phase lea
will become greater than 90° and tl
D.C. output of the phase comparator w
become negative. This will tend to t
duce the VCO frequency and lock will l
maintained with a slight increase ii
phase lead. Conversely, if the VCO fn
quency tends to decrease, the output (
the phase comparator wEl become pos
tive, which will tend to increase the VO
frequency.
It can be shown that the input signal ci
also lock to harmonics of the VCO fn
quency, or the VCO to harmonics of th
input signal (if the input signal is r
sinusoidal as previously assumed). It j
also possible to insert a frequency dividj
between the VCO and the phase cob
parator and by a combination of fn
quency divider and harmonic locking tl
ratio of VCO frequency to input frj
quency can be made to assume peculij
values such as 16/3 for example. Th]
opens up intriguing possibilities for frj
quency synthesis.
The capture process
Until now it has been assumed that tl
PLL is locked in. It is now necessary I
consider what happens when the circil
is switched on and the VCO is out |
lock, as it almost certainly will be. '
short answer is that the VCO hunts uni
it finds a frequency and phase to whif
it can lock.
Some understanding of the capture pJ
cess, as it is called, may fortunately j
acquired without mathematics if tl
behaviour of the circuit is examined I
certain points in the loop and cert if
assumptions are made.
To assist in the explanation, assume fid
that the connection between the Ll|
output and the VCO input is broken. 1
VCO, deprived of a control voltage, *
take up its free-running frequency whi
may be assumed to be lower than tl
input frequency. It has already beJ
assumed, when discussing the locked!
pll systems
elektor april 1975 - 413
Figure 1. A control system consists of an infor-
mation source Ax, a comparator circuit C, a
processing circuit B and a controllable quan-
FigureZ The elements of a PLL are: the phase
comparator 0, the low-pass filter LPF, and the
controllable oscillator VCO.
Figure 3. Showing how the output of the phase
comparator varies with the phase difference
between the input signal and the VCO.
Figure 4. Diagram to illustrate how the differ-
ence frequency waveform changes during one
cycle of the capture transient.
condition, that the VCO frequency in-
creases when the VCO control voltage
goes positive and decreases when it goes
negative. It may also be assumed that the
LPF completely removes frequencies
equal to the sum of the input and VCO I
frequencies, that it passes D.C. with no |
attenuation and that it passes the differ- [
ence frequency of the VCO and input [
signal with some attenuation, which de- I
creases as the difference frequency de-
creases (i.e. as the VCO frequency ap-
proaches the input frequency).
While the VCO is running free because of
the supposed broken connection a differ-
ence-frequency oscillation of constant
amplitude appears at the LPF output.
When the connection is re-made what
next happens must be examined careful-
ly. As pull-in has not yet taken place a
difference frequency still exists and an
oscillatory voltage is fed to the VCO con-
trol input.
Consider now one positive swing of the
VCO control voltage from trough to crest
(figure 4). The VCO control voltage is
going positive, therefore the VCO fre-
quency is increasing and the difference
frequency is decreasing. Because of the
decreasing difference frequency the
attenuation of the difference frequency
signal in the LPF will be progressively
reduced and the overall swing of the
VCO control voltage will have greater
amplitude than with the VCO free-
running. Figure 4a compares the positive-
going swings under controlled and free-
running conditions, starting from the
same trough potential and time. The crest
of the controlled swing is more positive
and it occurs later because the difference
frequency is decreasing.
Figure 4b shows what happens during a
negative (crest-to-trough) swing. Here the
VCO control voltage is going negative
and the VCO frequency is decreasing, so
the difference frequency is increasing.
Attenuation in the LPF is thus progress-
ively increasing; overall amplitude is less
than when free-running and the trough
occurs sooner.
Figure 4b is added onto 4a to show what
will happen during one complete trough-
414 — elektor april 1975
to-trough cycle of the difference signal.
The positive-going half cycle has a more
positive peak than the free-running dif-
ference signal. This ‘handicaps’ the nega-
tive-going half signal and its reduced
amplitude also helps to make the trough
more positive than it would be in the
free-running condition.
Later cycles of the capture-transient, as
it is called, cannot be compared with the
free-running waveform, but they follow
the same general pattern. Positive-going
swings have increased amplitude while
negative-going swings have reduced am-
plitude. This results in both crests and
troughs becoming progressively more
positive whilst the time interval between
them becomes longer. This means that
the VCO frequency will also increase
until a point is reached where one of
these swings of the control voltage
sweeps the VCO frequency through the
input frequency. More swings may occur
until the VCO has found the correct
phase relationship before lock-in actually
occurs.
Applications of Phase Locked
Loops
A PLL provides two information out-
puts. The VCO frequency, which is re-
lated to the input frequency, and the
VCO control voltage whose value de-
pends on the phase difference between
the input signal and the VCO output.
If the desired information contained in
the input signal is in the form of a fre-
quency change (i.e. frequency modu-
lation) then the PLL may be used as an
FM detector. Its advantages over ratio
detectors and coincidence detectors are:
less distortion, better suppression of in-
terference and the absence of LC circuits.
PLL’s are also useful in frequency syn-
thesis as figure 5 shows. In the example
given in figure 5a the condition for lock-
in is that fc/nv = f r and with a channel
spacing of Af we have Af = f r . The fre-
quencies delivered by the VCO are thus
multiples of the reference frequency and
it follows that the VCO frequency is
itself determined by the division ratio n v .
In many practical cases a variable-ratio
divider will not be able to accept a high
VCO frequency directly, so the VCO fre-
quency is fed first to a stable fixed-ratio
divider and from this to a stable adjust-
able divider. With this procedure it is
possible to divide down from a relatively
high carrier frequency to a low channel-
spacing frequency. This is useful in, for
example, aircraft VHF equipment.
In figure 5b an arrangement for fre-
quency synthesis is shown in which delta
pulses (needle pulses) recurring at the
reference frequency from a crystal oscil-
lator are fed into the phase comparator
together with the VCO signal. As delta
pulses contain the odd and even har-
monics of the fundamental frequency
the PLL can lock onto any harmonic.
Construction of a PLL
a. The VCO
Requirements for the VCO depend, in
the first instance, on the application of
Figure 5a. By inserting a variable-ratio f
quency divider between the VCO and the phi
comparator it is possible to obtain various f
quencies from the VCO using a single reference
frequency f r
Figure 5b. With this system a large number i
frequencies may be obtained by a simpk
method than in figure 5a, though at the ei
pense of stability which generally decreases i
Figure 6a. This VCO ci
good linearity and wi
up to 50 MHz.
Figure 6b. This VCO circuit consists of a
LC oscillator tuned and/or controlled by
varicap diode. If the oscillator is also used ft
tuning a receiver (i.e. as the local oscillator)
is known as a tuneable voltage-controlle
oscillator (TVCOI.
Figure 8. The symmetrical multiplier is used ^
almost all PLL IC's and can also be obtain
as an 1C in its own right. It may be co
structed successfully from disc re
components also.
used, provided that the low-pass filter can
vide sufficient suppression of the input
quencies. This type of circuit is used in
input section of an OTA and a PLL of
performance can, in fact, be built with
OTA type CA3080.
Figure 10. If RF transformers are used, a chea
multiplier may be built using four identic^
nrmnr
pll systems
elektor april 1975 — 415
Ire-
Ire-
1 of
tier
I
Illy
for
r) it
lied
the PLL. When it is to be used as an
FM detector the linearity (Frequency
change v. control voltage change) should
be as good as possible, while for fre-
quency synthesis this is unimportant but
high stability is essential.
Voltage-controlled multivibrators or
varicap-tuned LC oscillators, like those
shown in figures 6a and 6b respectively,
generally have to be made up from
discrete components, while integrated
PLL circuits, such as the Signetics 565
shown in figure 7, rely on the triggering
principle.
Where a PLL is to be operated with a
fluctuating supply voltage the VCO fre-
quency should be independent of volt-
age, or alternatively a stabilised supply
may be used.
b. Phase Comparator
The output from the phase comparator
or multiplier must be dependent solely
on the product of the signals fed into it.
This requirement is basically met by any
non-linear component, subject to the
proviso that the input signals also appear
in the output. It is important to ensure
that these signals have no detrimental
effect on the performance of the system.
An even more important requirement is
that the output should not contain any
D.C. components resulting from rectifi-
cation of the input signals, as this can
cause ‘mistracking’ and may even cause
the PLL to go out of lock.
If a balanced multiplier as shown in
figure 8 is used impairments such as these
can easily be avoided. The input signals
are suppressed by the circuit and no
rectification occurs. If suppression of
the input signals is not required it is
possible to use an asymmetric multiplier
such as the example in figure 9. A circuit
of this kind is included in the input of
an operational transconductance ampli-
fier (OTA) such as the CA 3080. This IC
performs well in PLL circuits.
It will be understood that rectification
of the input signals can occur in this case,
but nonetheless a satisfactory degree of
AM suppression may be achieved.
The best performance in this respect is
achieved when the VCO output is fed
into the asymmetric input and the input
signal into the symmetrical input. The
amplitudes of the signals should not
exceed 0.5 V and 0.05 V respectively.
The degree of AM suppression that may
be obtained is almost as high as with a
symmetrical multiplier.
If R.F. transformers are available it is
possible to use a diode ring modulator as
a multiplier as in figure 10, but this is a
rather old-fashioned method.
The simplest, but unfortunately also the
worst, solution for a phase comparator
consists of a single semiconductor device
that is fed with a VCO signal large enough
to switch it on and off continuously.
Because of the inevitable feedback from
the circuit to the VCO a buffer stage is
essential, as in the arrangement of fig-
ure 1 1 . The phase comparator here is
reduced to a mixer, so it appears that
any mixer may be used as a phase com-
parator. The problems that it introduces,
however, cannot be eliminated without
adjustment using expensive test equip-
ment. Symmetrical phase comparators,
on the other hand, give satisfactory re-
sults with very little outlay on test
equipment.
c. The low-pass filter
The low-pass filter (LPF) is the circuit
that determines the bandwidth of a PLL.
Simple RC filters, a few examples of
which are given in figure 12, usually
suffice. Examples b, c and d are suitable
for symmetrical phase comparators,
while a is applicable to asymmetric
arrangements. As a general rule resistor R
is already a component in the phase
comparator.
Although the calculation of component
values for the low-pass filter is easily
accomplished when using IC PLL’s by
referring to the manufacturer’s data,
sophisticated test equipment is needed
to evaluate the performance of a PLL at
frequencies in excess of 10 MHz. Filter d
is the most suitable for home-built
equipment.
The cut-off frequency of the RC combi-
nation formed by C 2 and the output
resistance of the phase comparator is
determined by the lowest frequency to
be detected (20 Hz in Hi-Fi FM). The
cut-off frequency of the second RC sec-
tion, formed by P(at its maximum value)
and C, both connected in parallel with
the output resistance, is determined by
the maximum PLL input frequency de-
viation. Any desired bandwidth, up to
a maximum determined by the loop gain
and the input signal amplitude, may now
be set with P.
Problems experienced with PLL's
Theoretically a PLL detector exhibits
great advantages over other FM detec-
tors, but in practice these are difficult
to realise fully. There are two basic criti-
cal factors:
1 . VCO frequency stability
2. Signal/noise ratio
416 - elektor april 1975
To obtain good stability the D.C. supply
to the VCO must be temperature-com-
pensated, and this applies also to the
phase comparator if the control input
to the VCO is asymmetric. In addition
the components whose values affect
VCO frequency should have zero tem-
perature coefficients. These requirements
are difficult to meet and in practice the
VCO centre frequency often drifts sev-
eral percent over the working tempera-
ture range. For this reason it is advisable
to choose the lowest possible working
frequency. The lowest usable working
frequency depends on the FM signal
bandwidth and with the 200 kHz usual
in FM broadcasting satisfactory opera-
tion is possible with a working frequency
as low as 450 kHz. Frequency drift at
this low working frequency may be
neglected; however, a receiver using this
principle must employ double conver-
sion techniques (i.e. it must be a double
superhet receiver) and will inevitably cost
more than a conventional receiver.
Both the VCO and the phase com-
parator generate some noise, so the de-
modulated signal level must be as high
as possible in relation to that noise. The
PLL output-signal amplitude is pro-
portional to the quotient of the devi-
ation f and the working frequency, which
in a receiver is of course the intermediate
frequency fjp. With an intermediate fre-
quency of 10.7 MHz and a deviation of
75 kHz this quotient is about 0.007,
while with an IF of 450 kHz it is 0. 17 so
that the lower frequency improves the
signal-to-noise-ratio by about 28 dB.
A PLL constructed from discrete com-
ponents, working at 450 kHz and using
the phase comparator of figure 8 and the
VCO of figure 6a, can achieve a signal-
to-noise-ratio of 60 dB on a stereophonic
broadcast.
Feedback PLL
As outlined above, the main problem
when using a conventional PLL as an FM
detector arises from the standardisation
on 10.7 MHz as an IF frequency. This
means that practically all commercially
available FM front-ends have an IF out-
put at this frequency. In addition, special
provision has to be made for the deri-
vation of an automatic frequency cor-
rection (AFC) control voltage from the
PLL. However, by removing some of the
components from the AFC loop in a
conventional tuner the local oscillator
can be used as a VCO. The linearity of
such a VCO can be quite good since the
75 kHz deviation is small in relation to
the working frequency (around
100 MHz). The reference frequency for
the phase comparator can be supplied
by a stable oscillator in which the fre-
quency-determining element is a quartz
crystal or a ceramic filter, so that VCO
phase jitter noise, which is relatively
strong at 10.7 MHz, is avoided.
Figure 1 3 is a block diagram of a feed-
back PLL. The aerial signal is mixed with
the output from the tuneable voltage-
controlled oscillator (TVCO) to give a
10.7 MHz signal that is fed through
Figure 11. This circuit may be used as a phase
comparator, but unwanted demodulation prod-
ucts arise due to modulation. This precludes its
use as an FM detector.
Figure 12. Of these low-pass filter circuits
version d is best for home construction as it is
least critical to set up.
Figure 11 Feedback PLL differs essentially
from conventional PLL insofar as it includes
the I F filter in the control loop. This results in
a substantial reduction in the IF signal devi-
ation, to the extent that the I F bandwidth can
be low enough for m to be unity or less. This
makes alignment of the bandpass filter and
component values in the lowpass filter ex-
ceedingly critical and for these reasons it is
better to choose a larger bandwidth. There are
a number of feedback PLL systems in which
the principal aim is to maintain the modulation
index as consistently as possible at unity. The
complexity of such systems, however, as well
as the difficulty of aligning them, limits their
use to radio amateurs with sufficient theoreti-
cal knowledge and to space-travel communi-
cation.
13
pil systems
decimal to bed converter
elektor april 1975 — 417
FI
Hits
lie
loll
.
an IF filter to the phase comparator. The
other input to the phase comparator
receives a high-stability 10.7 MHz refer-
ence signal from the reference oscillator,
thus, when the signal is locked in, the
TVCO follows the aerial signal deviation.
This means that the deviation of the
10.7 MHz signal is considerably reduced,
hence the name ‘Feedback PLL’. Because
of this reduced deviation the IF band-
width is much smaller than in a con-
ventional receiver.
In the article entitled ‘Modulation Sys-
tems’ the minimum bandwidth of an
FM signal is given as:
bmin. = 2(m+ l)f LFmax
and this relationship is valid when
m > 1. In a feedback PLL, however, the
IF-signal modulation index is consider-
ably less than 1 which accounts for the
reduced bandwidth. The significant
advantage of a feedback PLL system lies
in the IF bandwidth, which becomes
independent of deviation and in fact
depends only on the highest modulation
frequency. This gives improved signal-to-
noise ratio and lower distortion com-
pared to a conventional receiver, al-
though the degree of improvement de-
pends on the original modulation index
of the aerial signal.
For mono FM transmissions, with a
maximum modulation frequency of
15 kHz and a modulation index of 5, the
IF bandwidth in a conventional receiver
must be 1 80 kHz, whilst the bandwidth
in a feedback PLL receiver is only
30 kHz. The ratio is considerably less
unfavourable for stereo transmissions
however, as the highest modulation fre-
quency of 53 kHz means that the feed-
back PLL IF must have a bandwidth of
106 kHz. The principle of feedback PLL
was known before the introduction of
stereo FM broadcasting but unfortunate-
ly this did nothing to prevent the intro-
duction of multiplex stereo systems and
so any improvements that might have
been made in stereo reception were
thrown away.
It is still true to say, however, that a
feedback PLL receiver similar to fig-
ure 1 3 gives a considerable saving in cost
compared to a conventional receiver with
comparable performance. Feed-
back PLL systems are of particular in-
terest to radio amateurs, because signifi-
cant improvements in signal-to-noise
ratio may be realised if a low maximum
modulation frequency is specified. How-
ever, as far as the author is aware, little
work has been carried out in this field.
This is surprising as the principles in-
volved have been known for many years
and the VHF and UHF amateur bands
offer unlimited possibilities for experi-
mentation.
Summary
PLL is particularly suitable for frequency
synthesis and for demodulation of
FM signals. When used as an FM detector
the relative deviation of the input signal
should be as high as possible. This
involves the use of multiple frequency
conversion which is too expensive for
the consumer market and too compli-
cated for many home constructors.
Feedback PLL’s may be used at high
frequencies and offer the advantages of
reduced IF bandwidth and lower dis-
tortion with the absence of conventional
AFC. Full exploitation of the potential
of feedback PLL’s is probably too ex-
pensive for consumer applications. Never-
theless, simplified feedback PLL circuits
are feasible and are indeed cheaper than
conventional receivers. They should,
therefore, be of interest in consumer
electronics.
VHF and UHF radio amateurs are par-
ticularly well placed to take advantage
of feedback PLL techniques, as their
own experience makes them familiar
with the RF work involved.
A simple feedback PLL FM receiver will
be described in a future issue of Elektor.
M
J. Wittje
decimal to bed converter
This converter can be used as a manual
encoder which will convert decimal
coded signals into BCD codes and drive
digital circuits. Furthermore, the con-
verter can be used as a teaching aid for
explaining the BCD code.
One 1C and six germanium diodes are
sufficient for converting a decimal num-
ber into a BCD number. A switch for zero
is not provided because the converter
automatically indicates zero when all
switches are open. The reverse resistance
of the diodes must be as high as possible
(if necessary, check with an ohmmeter)
and the gate inputs can be provided with
a pull-up resistor connected to the posi-
tive supply voltage.
If the circuit is to be used to explain the
BCD code, the BCD-output conditions
can be indicated by means of LED’s. The
circuit for the required buffer stage is
shown in figure 2. H
418 - elektor april 1975
A. Seitz
fido
Fido is a new electronic game in
which an unfortunate dog is called
by four masters at the same time.
The command "Fido come" is given by means of a pushbutton. At each
push on one of the four buttons controlled by each player Fido jumps in
the required direction. However, the four masters and/or mistresses have
one handicap: After one successful command to Fido, the would-be Fido
owner who has given the order has nothing more to say for a certain
time. Then the other players can go on with Fido. If one of the players
succeeds in getting Fido into his kennel, the game is decided: Fido stays
where he is.
Construction and operation
l! Since Fido is clever enough to let him-
I self be represented by a small incan-
| descent lamp, he is not going to suffer
I from an otherwise unavoidable nervous
I breakdown. The worst that can happen is
I that after a prolonged fight for mastery
I over Fido our doggy will suffer from a
I flat battery.
I On the playing board nine lamps are
I arranged in a square (figure 1). On the
I extension of each side there is a lamp
I representing a kennel (so in total four).
| Furthermore, at each of the corners
| there are four push buttons with a pilot
lamp to indicate when a player can join
| the game. The buttons make Fido jump
i in four directions (away, towards, left or
I right with respect to the particular
| player). The photograph also shows that
the “gaming table” is provided with an
I on/off switch, an interval switch (coarse)
and an interval control (fine) for setting
I the obligatory rest period for the players,
j These switches can be calibrated “blood-
i hound/whippet” and “dog-tired . . . alert”
I respectively.
i Furthermore there is a switch to disable
the “rest” lamps and there is also a
starting switch. By pushing this button,
I Fido takes up his position in the centre
| of the field ; i.e. the middle lamp is alight.
By pushing one of his buttons, each
player can now try to direct Fido into
his kennel. Once a player has pushed a
button, he is obliged to take a breather
before he can push a button again. The
! lamps fitted near the buttons indicate
when the next command can be given.
Each player can give only one command
at a time. If an impatient player pushes
his button too soon, the penalty is a new
start of the waiting period. So Fido will
not respond to a command that comes
too early.
To make the game a bit more exciting,
the pilot lamps can be switched off, so
that each player must just guess when he
may next give a command.
by nine lamps arranged in a square. These
lamps are located at the intersections of
3x3 matrix rails. The signals for these
rails are driven by two left/right shift
registers. The clock pulses to the registers
are produced by the players pushing one
of the buttons. Since each player has
four buttons at his disposal, Fido can be
sent in all directions including the kennel
of another player.
The directing signals for left, right, up
and down are coupled into the registers
via the multiplexer. Once in a comer, the
dog can be made to jump into the kennel
situated below the comers as seen from
the player’s position. The register input
driving the “kennel” flipflop is so con-
nected that the command for jumping
is only followed if the other register, too,
is in the proper position. The lamp field
is blocked to prevent lamps from lighting
up after a jump into the kennel. At the
same time all register outputs are blocked
so that no more “kennel” flipflops can
be driven.
The game is started by pushing the j
starting button; then all the “kennel” I
flipflops are reset and the two shift I
registers take up a central position. In I
that case the middle lamp is alight.
The left/right shift register
Figure 3 shows how a flipflop can be!
turned into a “flipflopflap”. The inputs!
of each nand are connected to the out-l
puts of the other nands. Consequently,*
only one output at a time can be low!
(“0”). This “0”-signal produces a high!
output level (“1”) at all the other nands;!
these high levels in turn cause the low!
output level on the first nand. A negative-*
going pulse on one of the coupling rails!
causes all nands connected to this rail tol
change to “1”, whereas the nand whose!
output is connected to this rail ensures®
that this rail remains “0”.
If gates with a so-called totem-pole®
output are used (7400, 7420 and 7430W
the outputs must be separated by means!
a diode as otherwise none of the*
3
The block diagram
Fido’s position in the field is indicated
Figure 1. Artist's impression of Fido.
Figure 2. The block diagram. The command-
units also comprise the waiting time indication.
The push button "start" resets all "kennel"
flipflops and sets the registers at the central
positions, so that the lamp in the center of the
field lights up. Multiple connections between
the circuits are indicated by means of broad
Figure 3. The development of a multiple
flipflop starting from the fundamental prin-
ciple.
a. Two methods of drawing a simple flipflop
b. A 3-fold flipflop
c. A 5-fold flipflop.
„ j outputs would change to low (figure 3c).
Ll” With types with an open collector output
l? t I this is not strictly necessary, although
n I it is recommended to keep the input load
I of the pulse low.
I In that case the “0” must, after each
J pulse, shift one position to the right,
I left, top or bottom. So we need a
memory which remembers what coupling
rail is carrying a “0” signal before the
pulse, and a circuit that determines
in what direction the shift should take
place.
The memory is formed by Ci, (C 2 , C 3 ,
figure 4); the direction of shift is deter-
mined by two nands(N 3 and N 4 , N 5 and
N 6 , N 7 and N g ) which receive their sig-
nals via N] and N 2 . When the button is
pushed, say left, this is what happens:
Via Nj, connected as an inverter, the
“ 1 ” signal is fed to the nands N 3 , N s and
N 7 via the “left” conductor. At the same
time all the connecting rails are brought
to the “0” level via the diodes Di , D 3 ,
D 4 and D 5 . As a result, the nand N 9 ,
420 — elektor april 1975
fidt
Figure 4. Complete "shift register for a zero",
3-fold, for the matrix line of the horizontal
shift register. The vertical register is of the
same construction (description between
brackets).
Figure 5. Field with waiting time indication.
Depending on the type of field used, the trigger
unit is required several times. It serves to sup-
press contact bounce.
Figure 6. Diagram for Fido with nine lamps.
If the whole is fed from batteries, it is advisable
to supply the lamps from a separate battery
because pulses caused by switching (low
filament resistance of an extinguished lamp)
might interfere with the circuit. The bias of
Cg (figure 5) must also be obtained from a
separate battery because a maximum current
of about 200 mA can occur.
Nio or Nn , which has been at “0” level
so far, changes to “1”. Simultaneously,
a positive pulse is fed to the two adjacent
nands via the capacitor connected to this
output. The gate thus prepared by the
“1” signal via the conductor “left”
maintains the collecting line of its neigh-
bour at “0” until again via diode D] the
“0” signal disappears and the remaining
conductors become logically “1”.
The contact potentials of the diodes Di
and D 3 up to and including D s ensure
that the coupling rails reach the “1”
potential before the inputs of the gates 1
or 2. This is necessary to ensure that the
new main nand takes over the “0” signal
before the direction determining gate
changes back to “1”.
In the extreme positions for the “shift
register for a zero”, the “kennel”-
flipflops Ni2-N, 3 and N, 4 -Ni S are driven.
These may be driven only when the sec-
ond register reports the correct position.
The outer direction determining gates Nj
and N g , which drive the “kennel” flip-
flops require three inputs for that pur-
pose; one being coupled to the corre-
sponding matrix line of the other register.
Command-unit with indication
Figure 5 shows a command-unit with
four push buttons. The other units are
similar.
Via Pi and R 17 or Rig, respectively,
capacitor C 6 is negatively charged until
the voltage across C$ equals the sum of
the contact potentials of diode Di 0 and
the base-emitter junction of T 9 . The
latter is then conductive, so that Ti 0
causes the lamp to light up. The pilot
lamp indicates when a command can be
given. The waiting time can be adjusted
with Pi .
When pushing a button, say Si, Tj is
turned on by the negatively charged
capacitor C 6 , so that the emitter of T i
drops from +4,5 V to +0,7 V. This pulse
serves to drive the shift register.
Due to contact bounce, Fido is likely to
make wild and unpredictable jumps, or
just stays where he is. To avoid such
“disobedience”, each push button must
be connected to a trigger. Even the
shortest pulse at the base of Ti is suffi-
422 — elektor april 1975
fido
cient to cause the two transistors (T L and
T 2 ) to switch. As a result capacitor C 6 is
connected to the control line until the
voltage drop across Ri 3 caused by the
charge current is no longer high enough,
and the trigger returns to its initial
position. Then capacitor C 6 discharges
across R )7 (Rig) and P t .
The complete diagram
Owing to the large extent of the circuit,
some of the sections are represented as
blocks in figure 6. The positions indicated
by the coupling rails are represented by
“0”-signals. For the remainder, only
“1 ’’-signals are used; hence the inverters
7405 for inverting the signals. These
signals are fed to the lamp drivers 7440
which cause the lamps to light up when
all inputs are “1”.
Since only two of the four inputs of the
lamp drivers are used, all the others can
be connected to the positive of the
supply, which, however, is not necessary.
Once Fido has disappeared into a kennel,
that is to say. when a “0” signal has
reached the input of a goal flipflop, a
“ 1 ” is produced at the driver of the goal
lamp, and a “0” at the gate N 20 ,
which via the inverter N 2 | and six
diodes Du up to and including D !6 trans-
fers this signal to the outputs of the
inverters Ii up to and including I 6 . As a
result all the lamps in the field are ex-
tinguished. Furthermore, all the outer
direction-determining gates (figure 4) are
blocked (“0”-signal at the inputs that are
connected with the inverter outputs),
so that no further goal can be scored
by the now invisible Fido, if more but-
tons were pushed.
The start- or reset button returns the goal
flipflops and the registers to their initial
positions again. The middle coupling
rails must be connected to the reset
conductor via the diodes (D 9 in figure 4) .
The words “left”, “right”, “top”, “bot-
tom”, “vertical” and “horizontal” are
related to a group of push buttons which
is fixed by an arbitrary position of a
player and is called command-unit 1 . The
other command-units are numbered
clock-wise. The arrows in figure 5 are
related to the way in which Fido moves
as regards the player concerned.
Variations
The game can easily be changed. A first
possibility is to expand the field so that
the game will last longer (figure 7, accord-
ing to the principle in figure 3c). This will,
of course, increase the cost of the unit
by a considerable amount, especially if
the 25 lamp version of figure 7 is used.
Furthermore, it should be noted that the
field is in fact only suitable for four or
eight players, whereas the smaller field
can also be used by two without Fido
endlessly running up and down.
On the other hand, the field with
25 lamps can easily be connected to eight
command-units, so that eight “dog
lovers” can join the game.
A “mini Fido” is also a possibility if we
restrict ourselves to one register (see fig-
ure 3c), and if the “kennels” are placed
at the two ends of the row of lamps (fig-
ure 8). In spite of the simple set-up the
game can still be fun; playing with the
push buttons alone is most amusing. In
addition this version offers the possibility
of studying the register.
Of course, other possibilities can be
worked out, but then again it is up to the
reader to find an arrangement in accord-
ance with his taste and, lets face it,
budget.
M
elektor
services
to readers
EPS print service
Many elektor circuits are accompanie
by printed circuit designs. Some of thes
designs - but not all! - are also availabl
as ready-etched and predrilled boards,!
which can be ordered from our Canter-
bury office. A complete list of the)
available boards is published under the
heading ‘EPS print service’ in every issu<
Delivery time is approximately three
weeks.
As a further service, boards which are
taken off the regular service list at somi
future date will continue to be avaiHblt
in spite of this: delivery time will thei
be approximately six weeks. It shoult
be noted, however, that only board
which have at some time been publish
in the EPS list are available; the fact tha
a design for a board is published in
particular article does not necess
imply that it can be supplied by elekto
Technical queries
Members of the technical staff will
available to answer technical queri
(relating to articles published in elektor
by telephone on Mondays from 14.00
16.30. Queries will not normally
answered at other times.
Letters should be addressed to th;
department concerned: TQ = Technic
Queries. Although we feel that this is a
essential service to readers, we regret tha
certain restrictions are necessary:
1. Questions that are not related t
articles published in elektor cann-
be answered.
2. Questions concerning the connectio
of elektor designs to other units (e.
existing equipment) cannot normall"
be answered, owing to a lack of pract'
cal experience with those other unit'
An answer can only be based on
comparison of our design specifi
cations with those of the other equip-
ment.
3. Hieroglyphs or illegible handwritin
cannot be decoded: provided the
sender’s address is legible, the lette
is returned unanswered.
4. Questions about suppliers for com-
ponents are usually answered on the
basis of advertisements, and readei
can usually check these themselves.
5. As far as possible, answers will be on
standard reply forms.
We trust that our readers will understand!
the reasons for these restrictions. On the
one hand we feel that all technical
queries should be answered as quickly®
and completely as possible; on the othei
hand this must not lead to overloading]
of our technical staff as this could lead
to blown fuses and reduced quality
future issues ...
elektor april 1975
:
The block diagram of the interval switch
is given in figure 1 ; it consists of a pulse
generator, two monostable multivibra-
tors and a stabilizing circuit. A mech-
anism controls the automatic dia-
phragm and shutter of the camera.
The pulse generator consists of a UJT
(unijunction transistor) relaxation oscil-
lator with adjustable pulse recurrence
frequency. The output pulse drives two
interconnected monostable multivibra-
tors (MMVs) which control the mech-
anism for diaphragm adjustment and
H.U. Heinz A A
machine 16
Slowly-developing technological
processes or natural events cannot
be perceived because the eye is
generally not able to distinguish the separate stages. Such events and
processes can, however, be visualised by means of cinematographic time
compression. An interval switch linked with a camera enables it to make
single exposures at set intervals. When run at normal speed the film then
shows a process or event apparently developing continuously, but in a
much shorter time.
camera shutter. Because the circuit must
be suitable for battery supply, a stabil-
izing circuit ensures a constant voltage
throughout the battery life. Of course,
the circuit can also be fed from a mains
power supply.
MM V 1 controls the automatic diaphragm
of the camera. This diaphragm setting
is maintained until MMV2 operates the
shutter and resets the entire circuit to
its initial state.
The stabilizing section included a
battery voltage indicator which operates
Photograph 1. The time compressor system
for film cameras. The box mounted on the
camera contains the relays and the shutter
drive motor; the box beside it contains the
electronics.
with an ‘expanded scale’ and ‘suppressed
zero’ so that it only reads from about
12-20 V. Since the circuit will not
function correctly if the battery voltage
falls below 12 V, there is no point in
measuring below 12 V. It is simply
a waste of meter scale space.
The pulse generator
Figure 2a shows the principle of the
pulse generator. Capacitor Ci charges
via Ri to the breakdown voltage of the
UJT, to discharge again via resistor R 2
424 — elektor april 1975
I and the E-Bi junction of the UJT. The
I breakdown voltage of a UJT is an almost
fixed percentage of the supply voltage;
| usually between 60% and 85%, depend-
. ing on the type.
I * Positive pulses appear across resistor R2
, with a repetition frequency that can be
■ adjusted within certain limits by
changing R],
In the circuit of figure 2b, Pj is the
potentiometer with which the repetition
frequency is adjusted. The adjustment
range of Pj is determined by the series
connection of Ri and P2 in parallel
with Pj. Via the selector switch S2 this
combination is connected to the series
circuits R 3 + P 3 . . . R 6 + P 6 which are
connected to the supply.
I Terminal B2 of the UJT is connected
to the supply via resistor R 7 . This
resistor serves to reduce the temperature
dependence of the UJT.
I In the blocked condition, the E-Bi
junction of the UJT has a very high
resistance so that it is possible to
achieve relatively long pulse times with
large capacitances (220//) and high
resistances (maximum 1 M).
Switch S, j is combined with the on/off
switch; in the centre position C2 charges
rapidly via R 3 , so that the UJT can
I produce the first pulse the moment the
I on/off switch is operated. If the capaci-
| tor were not given an initial charge in
this way, the waiting time for the first
pulse would be 4 minutes in the worst
case.
Transistor T2 serves as an inverter, so
that the pulse generator supplies both
positive and negative pulses.
The Monostable Multivibrators
(MMVs)
j The two MMVs connected after the
I pulse generator are equipped with
1 thyristors with anode- and cathode-gates
I because these can fire on positive as
well as on negative pulses. Both MMVs
are of the same design, differing only
, in component values.
I Figure 3 shows the circuit of an MMV.
I Once thyristor Thj has been fired by
negative-going pulses on the anode-gate,
it remains on until the current drops
below the so-called holding current.
If in the anode circuit of the thyristor
a resistor is included of such a value
that the holding current of the thyristor
cannot be reached, the thyristor will
not fire.
If, however, a capacitor (C4) is now
| connected parallel to this resistor, the
thyristor will fire and the capacitor will
begin to charge. Since, however, the
charging current of a capacitor decreases
as the charge increases, there comes a
certain moment when the current
flowing through the parallel circuit of
resistor and capacitor drops below the
holding current, and the thyristor blocks
again. The capacitor then discharges
through the parallel resistor R 10
(figure 3).
A variable series resistance (P 7 + R n )
determines the charging time of the
capacitor and thus the time during which
the thyristor remains on. In addition,
this series resistance protects the
thyristor against excessive switch-on
currents. Via R 14 and D, the thyristor
drives switching transistor T 3 which
energises relay RLA. Diode D2 protects
the transistor against voltage surges when
the relay cuts out.
Current supply and measuring
circuit
The supply voltage is stabilized at about
1 1 V by ZDj and T s (figure 4). All
battery voltages can be measured under
loaded and no-load conditions via switch
S 4 . As long as the measured voltage
is higher than the zener voltage, a cur-
rent I flows through the parallel circuit
(R22 + P12); the resulting voltage drop
is measured with the measuring instru-
ment. The meter is adjusted to full-scale
deflection (f.s.d.) by means of P 12 . The
currents through the zener diodes
ZD2 . . . ZD 4 can be adjusted with the
potentiometers P 9 ... Pa - These zener
diodes ensure that only voltages higher
than the minimum voltages on which
the apparatus functions properly are
measured. The meter thus has a
‘suppressed zero’, i.e. it only reads from
(say) 12 V upwards since voltages below
this are of no interest. The whole meter
scale may then be calibrated for 1 2-20 V.
The residual battery charge can be esti-
mated on the basis of the difference in
meter deflections when readings are
taken with and without load.
The extra positions on S4 are for
testing other batteries in the camera.
The diodes ZD 3 and ZD4 can be chosen
to give a suitable ‘suppressed zero’
value for other battery voltages.
The complete circuit
The complete circuit given in figure 5
is intended for a Zeiss G.S-8 synchron-
ous camera. In this case the diaphragm
is adjusted by a motor, so that it remains
in the set position when the control
current is switched off. The camera is
fitted with two external connections for
electrically-operated remote release ; one
for single exposures and one for running
exposures. Before the release is operated,
the diaphragm must be properly
adjusted.
time machaj
The negative pulse produced at the
collector of T2 first starts MMV1 which,
via RLA1 (figure 6) switches on the
automatic aperture control for aboil
2 sec., giving ample time for this contra
to find its setting before the shuttei
opens. The moment MMV resets, i
positive pulse starting MMV2 occurs a
the anode-gate resistor (R13). As ;
result RLB is activated, closes contact*
RLB 1 , and starts a motor which drive
the camera shutter.
Although RLA is no longer energised
the diaphragm motor will hold the
aperture at its correct setting. The dia-
phragm drive can be switched off
altogether with S 3 , so that, for example;
an electronic flash can be used with 1
preset aperture. S 5 operates RLB
directly and can therefore be used foi
manual shutter operation.
There are almost as many automata
exposure devices as there are camen
types. Consequently the matching ol
the automatic operating equipment to
the camera diaphragm and shutte
mechanisms often calls for considerabl
Another type of automatic exposui
control which is found in most camera
nowadays uses a moving coil (as in
meter) to control the diaphragm accord
ing to the photocell response. In thi
case, the circuit operating the diaphragu
control must remain switched on whil
the shutter opens. This can be achieve
by providing an extra pair of contact
Figure 1. Block diagram of the time com
pressor.
Figure 2a. Circuit diagram of a pulse genet
ator using a UJT.
Figure 2b. Diagram of the pulse generator.
Figure 3. One of the two MMV’s with whie
the diaphragm control and shutter ar
operated.
Figure 4. This stage serves for voltage stabil
izing and checking the operating condition
of the batteries.
i i/nmin
is also required to function for these
shots, three components must be added
to the cathode gate circuit of Th 2 :
a 3n3 capacitor, a 470 12 resistor and
a diode (DUS). This must be done in
the same way as with MMV2 (here it is
Cs, D3 and Ris). The push-button of
figure 7 must then be connected direct
to the additional capacitor.
If the current consumed by the auto-
matic exposure control is known to be
small, the control can be left on con-
tinuously during time-compressed
filming. It will then be possible to
dispense with T 2 and associated
components, as well as with MM V 1 and
RLA. One pair of contacts on RLB
will suffice.
It can be gathered from what has been
said that adapting the circuit to a
particular make and model of camera
not only calls for a precise knowledge
of the camera; it also requires consider-
able experience in the field of elec-
tronics. Anyone who undertakes this 1
project should be capable of tackling |
Aligning the circuit
Before the apparatus can be used, the
following adjustments must be made.
1. Pi to zero, P3 to give maximum
pulse interval. This will be about
2 sec. for a mechanical shutter and
about 0.5 sec. for an electric shutter.
2. P4 , Ps , P6 to 1, 2, 3 minutes respect-
ively.
3. Pi in position ‘maximum’. Adjust P 2
until the difference between the
minimum and the maximum positions
of Pi corresponds to 1 minute.
4. P 7 to a time which enables the
automatic exposure control to re-
adjust by two stops.
5. Pg to give the minimum time the
shutter mechanism needs to operate
the shutter when the battery is low.
6. Adjust Sj and S4 to ‘off position,
P9, Pm and Pn to give 2.5 ... 5 mA
measured between the contacts of S4 .
Adjust Pi 2 to full-scale deflection
of the meter.
When choosing the zener dioi
(ZD 2 . . . ZD4) take into account
minimum voltages at which the equ
ment will still function properly at 1
temperatures. If the zener voltages
changed, other values may have to
chosen for the adjustment potent
meters.
Figure 6. Relay contacts for cameras
motor-driven or moving-coil diaphr
control.
nr iin
time machine
elektor april 1975 - 427
The shutter mechanism
(for mechanical shutters)
As is apparent from the previous
examples, cameras with electric shutters
are easily modified by bridging the
the release contacts by the relay
contacts. With mechanical shutters
however the release button must be
operated by a servo or other device.
No detailed data can be given on the
release mechanism because the con-
struction depends largely on the camera
used. The author used a Graupner
Varioprop-Servo from which the feed-
back potentiometer had been removed.
This was used to drive the shutter
release via a Bowden-cable type remote
release. Limit switches were incorporated
to limit the servo travel. A model control
servo which may be adapted to a shutter
drive for most cameras will be obtainable
in a shop for model builders.
Exposures with the time com-
pressor
To conclude with, some remarks about
the exposure technique. To ensure a
flowing motion, calculation of the
intervals should be based on 900 frames,
so that at a projection speed of 1 8 frames
per second the projection time is
50 seconds.
If the interval is indicated as t seconds
per frame (F), and the time in which
the compressed event takes place is T
hours, we have:
in which T is in hours, and t is in
seconds.
For an opening rose the interval for an
exposure time from 0530 to 2030
(exactly 1 5 hours) is
t = 4 x 1 5 = 60 seconds per frame.
When filming outdoors, don’t forget to
immobilize the flower in case it should
sway in the wind.
Editorial note
A number of notes as regards component
values may be made:
All electrolytic capacitors must be of the 16
or 25 V type.
For T 2 a BC 140 may be used instead of a
BFY 39. Furthermore, it is advisable to
connect a resistor of 1 k in series with the
base of T 2 .
In figure 4 transistor Ts (2N2219) may be
replaced by a BD 137 or BD 139. In many
cases this transistor will also have to be
cooled, certainly if the two relays draw con-
siderable current (over 100 mA).
Finally it should be noted that in figure 4
'+V b ' is the output of the stabilized supply.
So this point is the supply point ('©■) in
figures 5 and 7. The voltage is about 11V. M
marine diesel
Apart from ship sirens and fog horns,
builders of ship models are also interested
in imitating marine engine noises. With
only a few components the ‘marine diesel’
circuit lends realism to a model.
The noise produced by a diesel-driven
ship is made by the thump of the engine
and the regular puffing of gases escaping
through the exhaust. The noise of these
escaping gases is imitated by a small noise
generator in the circuit. The thump effect
is achieved by using an 1C in a trapezium
generator circuit, with the noise added
on the leading and trailing edges. The
figure shows the circuit. The base-emitter
junction of Ti is reverse biased to break-
down and the resulting noise signal is fed
to the non-inverting input of the opera-
tional amplifier. The feedback network,
formed by R 4 , R s , R 6 and C 3 then
determines the form of the trapezium
voltage. As long as the IC has not reached
saturation, the output produces a voltage
ramp with superimposed noise. The noise
is suppressed as soon as the IC reaches
saturation. An oscilloscope connected
to the output of the circuit should show
one of the waveforms drawn in the dia-
gram, depending on whether the DC-
connected or the AC-connected oscillo-
scope input is used.
If after completion of the circuit it is
found that the sound produced by the
model is too slow, certain modifications
may be made. Ci affects the noise; C 2 ,
Rs and R 7 determine the repetition rate.
The output of the circuit can be connec-
ted to the input of an amplifier. A resistor
(value to be found by experiment, de-
pending on amplifier sensitivity andinput
impedance) connected between the cir-
cuit and the amplifier prevents overdrive
of the amplifier. 14
noise
generator
Despite its simple design, this circuit is
a universal noise generator which pro-
duces a very high noise amplitude.
Transistor T! is connected as a zener
diode and is connected to the base of
the second transistor (T 2 ). The current
through the zener transistor, and hence
the amplitude of the noise, is adjusted
by resistor R, . This noise voltage is then
amplified by T 2 .
The supply voltage can be varied over a
wide range and, depending on the re-
quired output voltage, can be chosen
between 10 V and 30 V. At a number of
different supply voltages the following
noise output voltages were measured:
+V b = 12 V — 5 Vp. p
+ V b = 1 5 V — 8 V m
+ V b = 20 V — 10 Vp. p
+ V b = 25 V — 15 Vp- p
If required, transistor Ti serving as the
zener diode can, of course, be replaced
by a real zener of 6-8 V.
port 2
minidrum
The Minidrum described in the previous issue may, by the addition of
various extra circuits, be extended to a comprehensive manual drum kit.
Some new instruments, a three channel ruffle system and an automatic
bassdrum are described in this article.
The basic Minidrum contained only
three instruments, a bassdrum, a snare-
drum and a cymbal and so only three
channels of the TAP were used. Since
the TAP board has facilities for six
channels the design example given here
is based on six instruments. The number
of instruments may, of course, be ex-
tended to suit individual taste by adding
extra TAP boards, one for every six
additional instruments.
A pulse generator is included in the
design. This is intended to drive the
automatic bassdrum, but may be used
to drive other instruments either separ-
ately or simultaneously.
The ruffle system comprises three ruffle
channels driven by a single oscillator.
A pulse train appears at one of the
outputs when a finger is placed on the
appropriate touch contact. This may be
used to drive any of the instruments to
give drum rolls etc.
The first part of the Minidrum to be
described is the TAP circuit which
controls the instruments via touch con-
tacts.
The Minidrum TAP
Figure 1 is the circuit diagram of the
complete Minidrum TAP. It has six
touch inputs and six outputs, corre-
sponding to the six instruments used in
the design.
As described in the previous issue each
TAP channel consists of a COSMOS
inverter (1 j-I«) followed by a diode and
an integrating network. Hum from the
player’s skin causes the output of the
inverter to switch between logic 0 and 1 ,
charging capacitor (Cj-Cg). This output
voltage controls the relevant instrument.
The 47 k resistors (R 7 -Ri 2 ) limit the
base current of the one-shot associated
with each instrument.
Two types of RCA COSMOS IC may be
used for the TAP, CD4009AE or
CD4049AE. When using the former
diode Di must be included in the
circuit (see figure 1), if, however, the
CD4049AE is used, Di may be replaced
by a wire link on the p.c. board.
Due to the high noise immunity and
wide supply voltage tolerance of COS-
MOS circuits a sophisticated power
If if If
'MINIDRUM TAP'
SSS
430 — elektor april 1975
supply is not required, although some
form of simple stabilizer is desirable.
The Minidrum TAP p.c. board
Figure 2 is the p.c. board and component
layout for the TAP circuit of figure 1.
It is recommended that a socket be
used for the IC to avoid the possibility
of damage due to static or leakage from
unearthed soldering irons. A photograph
of the completed board is given in
figure 3.
The instruments
All the percussion instruments use the
gyrator board described in last month’s
issue, the circuit of which is given in
figure 4, but with component values to
suit the different types of instrument.
Table 2 gives the component values
which are common to all the gyrator
boards, while table 3 gives the compon-
ents which determine the characteristics
of the individual instruments. The
gyrator p.c. board and component lay-
outs are given in figures 5-13.
Each percussion instrument has two
inputs, input 1 and input 2. A mono-
stable multivibrator (one-shot) is con-
nected to each of these inputs and these
one-shots drive the gyrator. The output
from the TAP is used to drive input 1
while input 2 may be driven by the
ruffle system if desired. If ruffle is not
required on a particular instrument then
the monostable on input 2 may be
omitted, as in last month’s article.
As described in last month’s article, the
snaredrum has filtered noise mixed in
with the output of the gyrator. The
cymbal, brushes and maraccas are merely
filtered noise, with no gyrator input.
The p.c. board given in the previous
issue is used for the noise generator and
noise gating. If an instrument is to be
used with the ruffle system (the snare-
drum for example) then both gating
inputs of the snaredrum noise board are
used, one for the manual input and one
for the ruffle input. In the case of the
snaredrum these inputs are driven by
the one-shots on the gyrator board and
thus the one-shots on the noise board
may be omitted (figure 15). In the case
of purely noise instruments (cymbal,
brushes and maraccas) the manual TAP
drives the noise board directly and the
one-shot(s) must be used (figure 14).
If the ruffle system is not used then one
noise board will do for two instruments,
as was the case in last month’s article
where snaredrum and cymbal noise were
produced by the same board. The board
and component layouts are given in
figures 16-20. The component values
for the maraccas and cymbal noise
boards are given in tables 4 and 5, those
of the snaredrum noise board in table 6.
R x is added in the circuit for the
brushes. To mount this resistor on the
p.c.b., the connection Rm - C 2S is left
‘in the air’, and R x is connected between
this junction and the original connection
to T is (see figure 19).
The automatic bassdrum
Figure 21 is the circuit of the pulse
Figure 4. Circuit of the complete gyrator
board with two input monostables. The
component values are listed in table 3.
Figure 5. The gyrator p.c. board.
Figures 6-13. Component layouts for all the
gyrator instruments listed in table 3.
generator for the automatic bassdrum.
The circuit is desgined around an RCA
COSMOS IC, the CD4011AE, which is
a quadruple two-input NAND gate.
Gates Ni and N2 form an astable multi-
vibrator. Pulses from the output of Ni
are inverted and squared by Tj. C4 and
Rs differentiate these pulses and D2
clamps the output so that only positive
going pulses appear at the cathode of
D 3 . These pulses may be used to trigger
any of the instruments, but in the
system described here they are used to
drive only the bassdrum. The tempo of
the bassdrum may be adjusted from
about 40 to 240 beats per minute by
means of Pj.
In passing it may be noted that the
circuit of figure 4 may be used on its
own as a metronome, by reducing R3 to
15 k, R4 to 1 k and Rs to 4k7. C4 is
increased to 1 00 n and D3 is replaced
by a 47 S2 resistor. The circuit will then
drive a small loudspeaker directly, and
may be used as a self-contained unit
with a battery since power consump-
tion is quite low.
Instead of a mechanical start/stop switch
the automatic bassdrum of course uses
a TAP. N 3 and N4 are connected as a
set-reset flip-flop; touching the start
contact sets the flip-flop and touching
the stop contact resets it. In the reset
(stop) condition the output of N 3 holds
the inputs of N 2 high via Di and the
astable will not start. In the set (start)
condition the output of N 3 is low and
Di is reverse biased, so the astable runs.
The circuit is so designed that as soon
as the button is touched the circuit
produces its first output pulse, even at
low repetition rates. When the stop
button is touched the circuit stops
immediately.
Figures 22 and 23 give the p.c. board
and component layout for the bassdrum
pulse generator. Again it is recommended
that a socket be used for the IC.
Figure 24 shows a photograph of the
completed board.
The ruffle system
The circuit of the complete ruffle
432 - elektof april 1975
Figure 14. Noise circuitry for the Cymbal,
Brushes and Maraccas. See table 5 for the
values of the unmarked components.
Figure 15. Snaredrum noise circuit. Note the
absence of input monostables.
Table 4.
Components list for Cymbal, Maraccas
and Brushes, (figures 14 and 17-19) for
components common to all boards.
Where values differ see table 5.
Resistors:
R 42. R 53. R 58. R 60- R 70. R 75 = 10 k
R44. R45 . r 48- r 63- r 65- r 73 ■ 470 k
R 49- R 66 = 6k8
R 50' R 67 = 888 k
R 52- R 69 = 5k6
R 55 , R 72 = 27 k
R 56- R 76 = 100 k
R74.R77 = 270 k
R59 = 4k7
P 2 = 10 k preset
Capacitors:
C-)g = 100/1/10 V
Semiconductors:
Tl2' T 13- T 16- T 17.Tl8.
T21. T 22.T23 = T UN
Tl4.Tl5.Ti9 = TUP
D7. D 9.°10.Dl2.
d 13- d 14 = DUS
Table 5.
Components list for Cymbal, Maraccas
and Brushes, for components unique
to one particular instrument.
Cymbal
S
i
Brushes
R 43' R 61
10 k
10k
2k 7
27 k
10k
180 k
820 k
220 k
820 k
10 M
10 M
-
220 k
270 k
220 k
r 54
100 k
8k2
10 k
470 k
470 k
270 k
100k
8k 2
100k
R x
180 k
150 n
100 n
100 n
Cl8
68 n
120 n
39 n
C19.C26
12 n
lOOp
47 n
C20.C27
100 n
12 n
X
4n7
680 p
680 p
68 n
470 p
390 p
C23
4n7
470 p
390 p
C 2 4
150 n
100 n
1 A*
68 n
120 n
180 n
4n7
680 p
100 p
C29.C3O
100 p
680 p
100 p
10 n
10 n
2n7
D8.D11
DUS
DUS
— = wire
ink
Table 6.
Components list for the snaredrum
noise board (figures 15 and 20).
Resistors:
r 78- r 91 = 820 k
R 79- R 92. R 99 = 478 k
r 80. r 93 = 6k8
r 81> r 94 = 880 k
R 81a. R 94a= 18 M
R 82- R 95 = 188 k
r 83- r 96 = 5k6
r 84. r 88 r 97. r 101 = 18 k
r 85. r 98 = 15 k
R 86 = 4M7
r 87. r 102 = 188 k
R 89 = 4k7
Rl00. R 103 _278k
P3 = 10 k, preset
Capacitors:
C33 = 100 H. 10 V
c 34- c 39 = ®n2
c 35- c 40 = 22 n
C 36- C 37.C41.C42 = 2n7
C3S- 1n2
C43.C44 = 10 n
Semiconductors:
Di5,Di5,Di7,Di 8 ,Di 9 ,
D20'^21>^22 = DUS
t 24- t 25. t 27- t 28 = T UP
T2e.T29.T30.T31 = tun
' 1
■I
1 1
1
1 1
f
llif m T
1 1 r
r
r
r
elektor april 1975
Capacitors:
Ci = 100 H, 10
C 2 = 27 n
C 3 = 2/J2, 10 V
C 4 = 2n7
Semiconductors:
1C = CD401 1AE
Ti = TUN
D, = BAY 61, BA 220
D 2 ,D 3 = DUS
Figure 16. Noise p.c. board.
Figures 17-19. Component layouts for Cym-
bal, Maraccas and Brushes respectively.
Figure 20. Component layout for snaredrum
IC1=N1...N4=4011
Figures 22 and 23. The board and component
layout for the bassdrum pulse generator.
system is given in figure 25. The system
is very similar in operation to the
automatic bassdrum. i! and 1 2 form an
astable multivibrator and 1 3 serves to
buffer the output and improve the
waveshape of the astable. I 4 -I 6 form the
TAP control for the ruffle system. As
the three channels are identical only
one will be described.
When the touch contact is not being
touched the input of I 4 is held low via
The output is therefore high. C 4 is
charged via R 3 and current flows into
the base oPT 2 via D 2 and R 12 . T 2 is
driven into saturation so that the ruffle
signal appearing via D 6 is blocked. When
the contact is touched the output of I 4
switches at 50 Hz between ‘0’ and ‘1’
BD PULSE GEN.
436 — elektor april 1975
elektor april 1975 - 437
Components list
for figures 25 and 27.
Rl.R2. R 4. R 6- R 8 = 10 M
R3.R5.R7 = 47 k
Rg,RlO. R 12< R 13- R 15. R 16 = 10 k
Capacitors:
Ci = 220 /Li/1 0 V
C2.C3.C7.C8.C9 = 4n7
C4.C5.C6 = 1 *t 5/10 V (2*12/10 V
Semiconductors:
1C = CD 4009 AE or CD 4049 AE
Tf . . . T 3 = TUN
D-, . . . D 15 = DUS
Z x = Z-Diode (see text)
Input resistors for mixer preamp
(figures 29 and 30).
Figures 26 and 27. The p.c. board and
component layout for the ruffle system.
Figure 28. The completed ruffle board.
Figure 29. Circuit of the mixer-preamplifier.
boards given in this example is given at
the side of the figure.
It should be stressed again that this is
only an example and that any combi-
nation of instruments may be used to
suit personal taste. All that is required
is a little common sense and application
of a few simple rules. When choosing
the combination of instruments the
following points should be noted.
1. For each gyrator instrument one
gyrator board is required.
2. For the noise instruments (maraccas,
brushes and cymbal) one board will do
for two instruments, unless ruffle is
required, in which case one board is
required per instrument. If ruffle is not
used then the input monostables are
included on the board and the instru-
ments are driven direct from the TAP.
If ruffle is used then the monostable
on the input driven from the ruffle
board is omitted.
3. In the case of the snaredrum, if
ruffle is used then one noise board is
required for this instrument, both input
monostables being omitted and the
inputs driven from the ruffle board
Figure 31. Example of a complete drue
system with four gyrator instruments and twe
noise instruments.
Figures 32 and 33. Photographs of a comprt
hensive manual drumkit using all instrument
except brushes.
1
I
I
minidrum elektor april 1975 - 439
than clear Perspex as this will afford
some electrical screening.
In figure 32 three noise boards for the
cymbal, maraccas and snaredrum are on
the left. To the right of them are seven
gyrator boards and on the extreme right
the auto bassdrum. Along the bottom
of the photograph are the two TAP
boards and the ruffle board. The mixer-
preamplifier described in last month’s
article is at the top right-hand comer.
The circuit and board layout are given
in figures 29 and 30.
The mains transformer should be
mounted well away from the TAP and
ruffle boards and the mixer-preamplifier
to avoid hum pick-up. Note that 9 chan-
nels of the TAP or I'A boards are used
in this example.
The Minidrum will be on display (and work-
ing) together with many other Elektor pro-
jects at the 1975 London Electronic Com-
ponents Show at Olympia, May 13-16.
440 — elektor april 1975
compressor
£A Compressors are now being used
on an ever-increasing scale. They
may be found in tape recorders,
intercom systems and baby alarms,
public address systems, disco-
theques and of course broadcast transmitters. A compressor supplements
a manual volume control and allows a system to adjust itself to a wide
range of input signals with little distortion.
The design described here should find a wide range of applications with
the electronics enthusiast.
The aim of compression
Where signals with a wide dynamic range
| have to be processed it is desirable that
] as little distortion as possible should
LI occur. The designer of, say, a public
|j address system may have given much
I thought to achieving a good distortion
*1 figure, but this is of no avail if the
I system is overloaded by an enthusiastic
speaker shouting into the microphone.
j It is of course possible to prevent a
II circuit from being overloaded by
li attenuating the input signal with a
|l fixed or manually variable attenuator,
|l but then in the example above the
I person who mumbles into his notes
| would certainly not be heard.
| This is where a dynamic range com-
il pressor comes in. A compressor is
I basically an attenuator, or variable gain
II amplifier, which is controlled by the
If signal it is attenuating, either directly
|| or by a control voltage derived from the
|l signal. As the signal increases so does
I the degree of attenuation, so the com-
I pressor tries to keep the output signal
constant whatever the input. This can-
not be achieved in practice, but it is
possible to limit the output to a narrow
range over a wide range of input signals.
In a p.a. system (figure 1) a compressor
could be included between the micro-
phone preamp and the normal volume
control. The compressor, like death, is
a great leveller.
Compressor Transfer Functions
At first sight it would seem to be an
admirable aim to control the output
signal amplitude with the input signal
as in figure 2. This system has an
overall gain of - where K is a constant
and vj is the input voltage (for an
attenuator of course the gain is less than
1 ).
So V 0 = — = K.
vi
The output voltage is therefore constant
for all input voltages. This seems admi-
rable until one considers what happens
1
olume
'i — — lyf v c
Figure 2. A first approach to a transfer
function for a compressor. This is doomed
to failure however.
Figure 4. a. Voltage-current curve of a fila-
ment lamp. The resistance increases with
increased current.
b. Compressor using a lamp and a fixed
resistor.
c. T ransfer function of the compressor.
Figure 5. a. Voltage-current curve of a VDR.
b. Compressor using a VDR and a fixed
resistor, c. Transfer function of the com
Figure 6. Dynamic characteristics of variou
types of compressor in response to a sudden
burst of signal.
Figure 7. Block diagram of an active com
pressor using a peak detector to derive I
control voltage which alters the attenuator.
elektor april 1975 — 441
when vj is zero. The gain then becomes
infinite and this idea becomes unnat-
tractive.
A much better solution is to control the
output signal with the output signal,
which at first sight may seem odd.
In figure 3 however it can be seen that
the gain is — .
v 0
, Kvj
Therefore vo = — .
2 v °
or vo = Kvj
This is a square-law compressor func-
tion. Of course, other functions may be
achieved, notably logarithmic, where
vo = K log vj.
Practical Compressor Circuits
There arc many different kinds of com-
pressor circuit. One of the oldest and
simplest circuits makes use of the non-
linear resistance of an incandescent
lamp, whose resistance increases as the
current through the filament increases.
In figure 4 the resistance of the lamp,
which forms the upper limb of the
attenuator, is low at low signal levels so
only a small portion of the signal voltage
is dropped across it. At higher signal
levels the resistance increases and a
larger proportion of the signal voltage
is dropped across the lamp. The output
signal therefore does not increase as
much as it would with a normal attenu-
ator. The thermal inertia of the lamp
filament means that this circuit cannot
follow the actual signal waveform but
only the envelope (provided the fre-
quency is not too low) so distortion
produced by the circuit is fairly small.
The thermal inertia of the filament
means, however, that the circuit cannot
respond quickly to sudden increases in
signal, so that associated circuitry may
be overloaded whilst the lamp resistance
is changing. Also the range of this type
of compressor is limited.
An alternative solution would seem to
be the use of a voltage-dependent
resistor (VDR) as in figure 5. This has
a voltage versus current curve which is
approximately the inverse of that of the
lamp, so it is included in the lower limb
of the attenuator. As the signal is
increased the resistance of the VDR
decreases so a smaller proportion of the
signal appears across it. The response
time of a VDR is quite fast so that it
will follow sudden increases in signal
amplitude, but unfortunately it can
also follow the signal waveform so that
instead of compressing the envelope
amplitude whilst preserving the wave-
shape it simply ‘rounds off the signal
peaks thus introducing distortion. None-
theless, in certain applications where
distortion can be tolerated, such as
amateur radio transmitters or intercoms,
it does have its uses.
It thus appears that the compressor
designer is caught between two stools.
A slow-acting device will cause little
distortion on sustained large signals,
but will not react sufficiently quickly
to prevent momentary overloads of the
equipment, whereas a fast-acting com-
pressor will react in time to prevent
overload, but will of itself introduce
distortion. Here, however, an unusual
aural phenomenon comes to the de-
signer’s aid. The ear is incapable of
detecting even large amounts of distor-
tion in transients, so that if a fast-
acting compressor is applied to a sudden
increase in signal it will prevent gross
overloading of the system whilst the
distortion it introduces will be unno-
ticed. Once the compressor has limited
the signal, however, the ear can detect
the distortion it introduces, so on
sustained loud passages the slow re-
sponse of the lamp-type compressor is
required. In fact what is required is a
compressor with a fast attack and slow
decay characteristic.
442 — elektor april 1975
The characteristics of various types of directly by the signal on which they
compressor are given in figure 6. The operate, but for a device with different
triangular waveform was used to show attack and decay time constants it is
how distortion is caused by a fast- necessary to turn to active circuits. In
acting compressor. figure 7 the signal passes through the
The discussion has so far been confined input stage and into a voltage-controlled
to passive devices that are controlled attenuator. The output voltage is taken
parts list:
Cl2.Cl3 = 47#i. 10 V
Rl,R4,RlO,Rl2 = 10 k
Ti,T 3 to Tg = BC 109C
Di = zener diode 2,7 V
T 2 = BC 179C
D 2 to D5 - germanium diode
R 3 = 4k7
R 5 = 220 fi
matched pairs A A 1 19
Dg to Dg - silicon diode 1N914
R6- R 17. R 20. R 26 = 22 k
capacitors:
R 7 = 1 k
Ci = 100 n
R 8- R 15- R 16 “ 330 k
C2.Cn = 1 n. 10 V
Rn = 270 k
C 3 = 180 p
R 13- R 14- R 25 = 3k3
C 4 - 100 H, 16 V
for V b - 9 Volt: R 18 ,R 19 = 270 £2
R 24 “ 47 k
C5,Cg,Cio “ 560 n
and R 23 = 1k8
R 27 = 120k
C 6 = 100p, 4 V
for V b = 12 Volt: Rl8.Rl9 = 330£2
Pi = preset 22 k
C 7 .C 8 = 2,2 H, 10 V
and R 23 = 1k5
Figure 8. An LDR used in a voltage-controlled
attenuator. This circuit suffers from slow
response due to the inertia of the lamp and
LDR.
Figure 9. An r.f. carrier type of compressor.
The filter eliminates harmonic distortion of
the carrier caused by the attenuator and also
eliminates control-voltage noise.
Figure 10. Voltage-current curve of a diode
and circuit of a simple diode attenuator.
Figure 11. Balanced type of diode attenuator
eliminates control-voltage noise which appears
in common mode.
Figure 12. The circuit of the final compressor
design.
Figure 13. The printed circuit board and
component layout of the compressor.
elektor april 1975 - 443
from the output of the attenuator and
is also fed to a peak detector which
rectifies the signal. The rectified voltage
charges up the capacitor C via the
potential divider consisting of Rj and
R 2 . The time constant is
The voltage on C increases the attenu-
ation of the voltage-controlled attenu-
ator as the signal increases. If Rj is small
C charges up quickly but since the dis-
charge path for C is via R 2 only, the
decay time constant can be made as
large as desired so that the voltage on
C will not follow the signal waveform.
as otherwise the variations in control
voltage with varying signal levels will
appear as spurious noise at the output.
One way of achieving this would be by
using a light-dependent resistor (LDR)
as the lower limb of the attenuator, as
in figure 8. This would be controlled by
a lamp driven from the control voltage.
Unfortunately problems arise due to the
slow response of both the lamp and the
LDR. Another rather elegant solution
is to amplitude-modulate the signal onto
a carrier and to vary the modulation
depth by a voltage-controlled amplifier
stage (figure 9). The compressed modu-
lated signal is then filtered to remove
control voltage noise and distortion
(mainly second harmonic) and is then
demodulated, resulting in a ‘clean’
compressed signal. Intermodulation
distortion can still occur, but this can
be minimised by proper circuit design.
The design chosen for the final circuit
to be described was a diode attenuator.
In its simplest form (figure 10) it suffers
from two disadvantages.
1. The signal voltage will itself vary the
attenuation as with a VDR thus causing
distortion.
2. The control voltage will appear at the
output superimposed on the signal thus
producing spurious noise.
The first problem may be overcome by
making the signal small compared with
the control voltage so that it has little
effect. The second may be prevented by
using a balanced attenuator of four
diodes as in figure 1 1 . The signal appears
differentially at the input of the differ-
ential amplifier and is therefore ampli-
fied. The control voltage, however,
appears in common mode and is there-
fore rejected.
The voltage-controlled attenuator
Whilst the derivation of a control voltage
from the signal is a relatively simple
matter the design of a suitable voltage-
controlled attenuator is another matter.
Ideally the attenuator should be electri-
cally isolated from the control voltage
The Final Circuit
Figure 12 shows the circuit of
l
444 — elektor april 1975
disc preamp
Compressor Specrficatio
Input impedance
180 k
Output impedance
25 k (do not
load with less
than 100 k)
minimum
Gain with Pi at
60 (max. input
voltage = 1 V)
maximum
Maximum (compressed)
150 (max.
input voltage =
30 mV)
output voltage
Maximum distortion
(gain ■ 60)
a. below compression
500 mV
threshold
b. at maximum (1 V)
0,4%
Maximum control
current through diode
5%
bridge
350 pA
Power consumption
10 mA at 9 V
disc
preamp
A preamplifier-equaliser for
magnetic pickup cartridges has to
meet quite exacting requirements.
Values for gain, noise level and maximum input voltage which will
guarantee trouble-free operation under all conditions are not so easy to
achieve. The well-known two-transistor configuration, operating from a
12 ... 18 V supply, invariably falls short on gain and overdrive-margin -
unless it is designed for a low nominal output voltage (about 30 mV).
An alternative approach is to make use of a good integrated amplifier.
The design about to be described, which meets all the requirements,
employs a SN 76131. An almost identical I.C. is the pA 739.
compressor intended principally for
speech applications. The circuit has an
input stage with adjustable gain which
is sensitive enough to be driven by a
magnetic microphone. This is followed
by a phase splitter which produces two
antiphase signals to feed into the
differential stage T4, T 5 . The com-
pressed output is taken from the
collector of T4 which should not be
loaded with anything less than 1 00 k
as this would upset the circuit oper-
ation. A class B-type stage T7, Tg drives
the peak detector Dg , Cu . The control
voltage appearing on Cu is buffered by
the emitter follower T« and is fed to
the diode bridge D 2 ... D s . Di is a
threshold control which determines the
point at which compression starts. is
simply a constant current source for the
'j differential pair.
1 1 The board and component layout for
I the compressor are given in figure 13
j and the performance figures in the table.
I I At first sight it may seem that the
I distortion with the compressor oper-
[I ating is rather high but compared with
the distortion when an amplifier is over-
I loaded it is minimal.
Applications of the compressor
J This compressor is sure to find a whole
J host of applications. It can be used to
1 control the recording level in a tape
L- recorder to prevent overloading of the
j tape. It can be used in amateur radio
I installations to achieve the largest
|| possible modulation without over-
modulating so that maximum range can
be achieved. It can be used in a car
radio so that quiet passages may be
heard above the engine noise without
loud passages being unbearable. The
I range of applications is limited only by
’ the ingenuity of the constructor -
remember, a compressor rules the waves
I I (somewhat straighter than they were
f originally!).
Bibliography:
Electronic Engineering, January 1973.
I Radio Elektronika, January 1959.
To make optimum use of the possi-
bilities for groove-modulation, gramo-
phone records are cut with low audio
frequencies attenuated and high audio
frequencies boosted (with respect to
1 kHz). To simplify playback equalis-
ation, a single weighting curve has
been standardised throughout the
world - the IEC disc-cutting character-
istic. (This curve originated as the RIAA
standard: Record Industry Association
of America).
The disc-cutting engineer arranges for a
‘0 dB standard (reference)level’ in the
taped programme to produce a stylus
tip-velocity about 14 dB below the ‘safe’
drive-level, to provide headroom for
instantaneous signal peaks. 0 dB standard
level (corresponding roughly to the
average level in loud passages) is typi-
cally 39 mm/sec tip peak velocity at
1 kHz. Standard level on carrier-channel
discs (CD4 and UD4) is lower, about
22 mm/s.
Experience indicates that wide-band
cartridges suitable for carrier discs
deliver 70 ... 140 pW for each mm/sec
of tip velocity. The usual ‘hifi’ cartridges
deliver about 6 dB more. (Note that
sensitivity specifications are usually
given in RMS millivolts per peak centi-
metre per second). So the input to the
preamplifier at standard level 1 KHz will
be about 1 ... 10 mV peak.
What are the consequences of all this for
the preamplifier?
Suppose it is the intention that the
output voltage at standard level be about
100 mV RMS with the lowest-output
cartridge. The closed-loop gain must
therefore be 100 at 1 KHz. Now allow
20 dB of extra gain for IEC equalisation
at the lowest frequencies, not including
20 dB of negative feedback (which
should reasonably be maintained at the
‘low end’). This tots up to an open-loop
gain of at least 80 dB! Ten thousand
times. That seems to eliminate the two-
transistor configuration.
The SN 76131 integrated circuit, with
the chosen lag compensation, has a
typical open-loop response according to
the upper dashed curve in figure 1 . The
Figure 1. The desired closed-loop gain curve
follows the IEC (RIAA) disc equalisation
characteristic, with a mid-band gain (1 KHzl
of 40 dB (heavy line). The open-loop gain
must be at least 20 dB greater; the SN 76131.
with the chosen lag compensation, provides
this with a margin of about 10 dB (upper
dashed curve).
Figure 2. The heavy line is an estimated
contour for the highest voltage delivered to
the preamplifier by a high-output dynamic
cartridge. The preamplifier cannot be over
driven by the highest input voltage; the upper
dashed line is the overdrive threshold for the
disc-preamplifier with SN 76131. This cl-
the maximum-input contour by approximat
10 dB.
Figure 3. The maximum RMS output l~
produced by the preamp when used with
high-output cartridge follows the thick ;
tour. The dashed line indicates the maxim
output capability. The safety margin is h
once again about 10 dB.
etektor april 1975 - 445
lower dashed curve indicates the mini-
mum requirement (80 dB at the low
end, reducing as the closed-loop gain
- i.e. the bold line in figure 1 - falls
according to the IEC curve). The con-
clusion is that there is about 10 dB of
open-loop gain to spare at all fre-
quencies, which will accomodate 1C-
tolerances etc.
Overdriving the input
To find the maximum input voltage
which can occur, one must start with
the highest-output cartridge. This will
deliver, as shown earlier, about
5 . . . 10 mV peak at standard level.
The maximum level encountered on the
disc is nominally +14 dB relative to
standard level. This indicates a nominal
maximum input voltage of 25 . . . 50 mV.
(At 1 KHz of course). It is clearly
advisable to regard this figure, with due
respect, as nominal. One might encoun-
ter a cartridge with still higher output
or some disc manufacturer may fully
exploit tracing-compensation, to cut a
clean signal at more than +14 dB . . .
The absolute limit (set by ‘slope-over-
load’ at the inner radius of LP discs) is
presently about 350 mm/s (+18 dB) -
but a 33 disc also has outer grooves and
they can be cut at a level 6 dB higher.
This means that in theory the
maximum output level for the highest
output cartridge is about 200 mV! With
the circuit arrangement given, the
SN 76131 will accept 80 mV at the
input (thick dashed line in figure 2).
The same figure can be used to estimate
the effect of amplifier noise. The wide-
band noise level, referred to the
SN 76131 input, is 2 /iV (RMS). This is
—68 dB in the figure (0 dB = 5 mV
RMS). For the least sensitive cartridge,
this noise level is -54 dB relative to
standard level for CD-4 or UD-4 discs.
Assuming maximum signal level to be
+14 dB the overall S/N ratio is (for this
worst case) 68 dB. Manufacturers esti-
mate that the S/N ratio possible with a
first-rate LP pressing is about 70 dB.
Conclusion: pass.
Figure 2 can be used once more to
determine the hum-level requirements.
The IEC bass-lift now aggravates matters:
to achieve a hum level 60 dB below
standard level, with a fairly high-output
cartridge (5 mV RMS at 1 KHz), it
becomes necessary to keep the hum
voltage at the input below 1 pV! This
can be achieved, in general, by providing
good screening for the input circuit and
for the preamplifier itself (signal-return
inside the cable-screen, the latter bonded
to signal-earth at the amplifier end
only), and by properly smoothing
(preferably regulating) the DC supply.
The sensitivity of the SN 76131 to
interference on the DC supply rail is
quoted — under operating conditions
rather different to the above - as
50 mV/V. (i.e. 50 /rV apparent input for
each volt of supply disturbance). To
achieve the 1 £iV hum level just men-
tioned means keeping supply ripple
below 20 mV. A simple active circuit
will readily meet this requirement;
446 — elektor april 1975
disc preamp
simple smoothing of a ‘raw’ DC supply
would probably be inadequate or too
expensive (or both!).
Clipping at the output
The requirement that the input circuit
is not overdriven will not by itself
quarantee that the amplifier as a whole
operates within limits. The output
circuit can still ‘run out of voltage or
current swing.
Taking the combination of a sensitive car-
tridge and the maximum disc modulation
likely to be encountered, one can esti-
mate the highest level of output signal
that the preamplifier will have to
deliver. This can be done by combining
the closed-loop gain characteristic
(figure 1 , thick line) with the maximum
cartridge output contour (thick line in
figure 2). The result is shown in figure 3
(thick line). The conclusion is that the
voltage swing at the output can be as
high as 2.5 V RMS (7 V p-p).
The clipping level for the SN 76131
depends on the supply voltage and on
the load impedance. The case of V + = 30
and Rl = 10 K, where the 1C can deliver
about 7 V RMS, is shown dashed in
figure 3. This reserve should take care of
all eventualities. If one considers a brink-
of-disaster capability of 3 V RMS, then
the combinations 18 V/5 K, 14 V/10 K
and even V+ = 12 (at Rl = 50 K) are in
order. Even under these conditions,
current clipping due to the load of the
feedback network on the output (at the
highest audio frequencies) and slew-rate
limiting (due to the early open-loop
rolloff) are not expected to occur.
Integrated circuit
The circuit was designed around the
specified SN 76131 by Texas Instru-
Figure 4. The pinning of the IC's SN 76131,
TBA 231, TCA 590C, [lA 739C and LM 1303
(figure 4a) it identical. The internal circuit
diagram (figure 4b) however only applies to
the SN 76131.
Figure 5. The circuit diagram of the equali-
preamplifier. An integrated voltage regulator,
when required, can be connected between the
points A and B (see text).
Figure 6. PC board and component layo
for the equaliser-preamplifier. All exte~
connections are made to one edge of the PC5
board, so that it can be used as plu*
module in a complete control amplifier.
Figure 7. Illustration of the preamplifier bo-
as plug-in module.
fcc preamp
Table 1. The most important specifications o
the SN 76131 and / JA 739C.
II r™-
36 V
±5 V
500 mW
1 V nilt swinq
1 ... 26 V*
Open loop gain typ
18000*
Open loop gain min
6500*
z in *VP-
150 KS2*
Z in min.
37 Kf2*
Z out (1 KHz)
Crosstalk (10 KHz)
-140 dB*
1 * These values apply for
R|_ = 50Kft
V+ = 30 V;
r merits. According to the maker’s data
[sheets, the Fairchild pA 739C and the
TBA 23 1 are almost identical and
perform well in the circuit. The
IC’s are pin compatible (see
4a). Two other IC’s with the same
are the Philips TCA 590 C and
Table 2. Main specifications of the disc pre-
amplifier described here.
the LM 1303 by National Semiconduc-
tors. This last device has lower specifi-
cations for gain, noise and drive level -
it will probably work acceptably in the
preamplifier, but we have not checked
this.
The internal circuit of the SN 76131
(and the TBA 23 1 ) is given in figure 4b.
Except for the output transistor, the
M 739C is identical. Table 1 lists the
most important characteristics of the
device. The TCA 590C has an additional
class B output stage, while the LM 1303
circuit dispenses with the stabilising
diodes and with the current sinks for
the second long-tail pairs.
The external circuit
Figure 5 gives the complete circuit dia-
gram of the equaliser-preamplifier. The
open-loop response is set up by C 4 /C s /R 3
and C11/C12/R11; it follows the appro-
priate dashed curve in figure 1. The IEC
correction networks are R1/R2/R4/C1/C3
and R12/R13/R14/C13/C15. R s and R] 0
take care of the DC biassing. With the
values given, the correction obtained
using 5% components is within 1 dB of
the IEC (RIAA) standard.
The input blocking capacitors C 7 and
C9 should not be replaced by larger
values or by electrolytics. This could
lead to undesirable switch-on phenomena
(‘plop’ or even momentary oscillation).
The values given will not affect the bass
response (which is 1 dB down at 20 Hz).
It has already been pointed out that the
supply ripple must be well filtered.
A typical regulated supply will meet
the requirements, but a ‘raw’ supply
followed by resistor-electrolytic filter
will usually cause too much hum. In this
case one can use an IC voltage regulator
which will deliver 24 ... 30 V at 15 mA
(or more), e.g. the Fairchild jiA 78M24HC.
The printed circuit board (figure 6) has
a position for this regulator. If such a
device is not to be used, the points A
and B should be bridged.
To simplify assembly, all external con-
nections have been placed at one edge
of the PC board, using standard grid-
spacing. A control amplifier which will
be published at a later date has a PC
board designed to accomodate the disc
preamplifier as a plug-in module
(figure 7).
Table 2, in conclusion, summarises the
most important specifications of the
equaliser-preamplifier for disc records.
Lit.: Texas Instruments data sheets for
SN 76131.
elektor april 1975
a/d converter
The convert a voltage
C^IIIhkSI ■ B5S to a frequency such that the
V frequency is accurately
proportional to the voltage is one which arises in many different
electronic systems. Some digital voltmeters use this principle. The
voltage to be measured is converted to a proportionate frequency, which
is then measured by a conventional counter circuit, and the result
displayed digitally. In other cases, the requirement is to have a reading of a voltage existing some distance
away. In this case the long cables, with their appreciable DC resistance, produce a voltage drop if any current
is taken by the measuring instrument, and errors result. If however the information is carried over the cables
as a frequency, although the amplitude may fall the frequency will not change. Increasing use of digital
computors, digital logic IC's, digital displays, etc., produces many more applications.
A previous design for a convertor circuit gave reasonable performance. However, further work produced
several relatively minor changes which improved both linearity and temperature stability, resulting in the
circuit described below.
It is relatively difficult to convert voltage
to frequency in a direct manner, if good
linearity is to be maintained. However,
the reverse operation, frequency to volt-
age, is much easier. With this in mind,
the method used here is firstly to convert
voltage to frequency in a circuit which in
isolation would not be very linear. The
output frequency is however then con-
verted back to voltage in another circuit
(which this time is highly linear) and the
output voltage used in a negative feed-
back loop path so as to linearise the
whole system. The overall linearity of
the system will then approach that of
the frequency-voltage convertor, pro-
vided the feedback loop gain is high.
Figure 1 shows the block diagram. The
high gain differential comparator (A)
accepts the input voltage and compares
1
frequency/
differential
amplifier
u \®
voltage/
* rTn
1
-o
li
. /
© |
it with the feedback voltage. The voltage-
frequency convertor (B), which can be
relatively non-linear, is driven from (AX
Its output provides the system output,
and also drives the highly linear fre-
quency-voltage feedback stage (C). j
The Basic Circuit
This circuit is shown in figure 2. A*
IC type 741 is used in conventioni
manner as the differential comparator
The system input voltage is applied U
the non-inverting input pin 3, and thf
feedback voltage to pin 2. The outpu
from pin 6 then regulates the frequenq
of the next stage in such a way that th«
two inputs remain almost identic^
Capacitor Ci provides AC negative feed
back, to prevent appreciable AC sigr
appearing at pin 6.
The IC type 709, with the other cc
ponents in the dotted line box (B) afl
figure 2, together form a square wav
oscillator. Consider first the case wheJ
Q'” all
©_
a/d converter
elektor april 1975 - 449
Figure 1. Block diagram of the complete
Figure 2. The basic circuit.
Figure 3. Showing extra transistor used to
improve zero error.
Figure 4. The final circuit of the convertor.
The conversion factor is set by Rig.
there is a negative voltage on C 3 . Of the
two differential inputs of the 709 (which
is a differential amplifier), pin 2 will be
more negative than pin 3, due to current
in R 6 from the negative rail. The output
at pin 6 will therefore be hard positive,
holding T 2 in saturation. Provided the
741 output is sufficiently positive, C 3 will
charge up via R 3 until it raises pin 2
of the 709 to slightly above pin 3. As
soon as this happens, the 709 output
pin 6 rapidly goes negative, thus cutting
off T 2 and allowing the system output
to rise to a voltage determined at about
8.2 V by D 6 . Immediately, the voltage
pin 2 is drawn even more positive by
current through R 7 . Thus the action is
regenerative. This condition now remains
for a time 1 1 , which is determined by the
values of R 7 , Rg, R 9 , Rio, and C 4 . (The
value of ti is not affected by the voltage
on C 3 , because as soon as 709 pin 6 goes
negative, C 3 is driven rapidly negative
via R 14 , D 4 , and Ri S .) C 4 is charged
positively via R 9 and Ri 0 until pin 3
becomes more positive than pin 2, at
which point the 709 output reverts, T 2 is
once more turned on, and the cycle
starts again with C 3 charging up. The
result is a series of rectangular pulses at
the output, whose width is ti and whose
amplitude is constant (at the Zener
voltage). Their PRF will however be
determined by the time taken to charge
C 3 , and hence by the 74 1 output voltage,
so that overall the 741 input voltage con-
trols frequency. D s is included to protect
T 2 from excessive reverse voltage on its
base. The circuit including Ti , R u , R 12 ,
Ri 3 , D 3 , and C s is put in to discharge C 4
to zero at the end of the period t, , and
is driven by the positive-going step
change from the 709 pin 6. The regen-
erative action, via T 2 , is speeded up by
capacitor C6 .
The frequency-voltage convertor is sur-
prisingly simple, comprising only R 4 and
C 2 ! It merely smooths out the AC com-
ponent of the rectangular wave, leaving
on C 2 the DC component, whose value
is exactly proportional to the PRF, or
frequency.
Improved Performance
It is a defect of the above system that
the saturation voltage of T 2 (i.e. its
collector-emitter voltage when turned
hard on) is not exactly zero, but can be
something around 40 mV. Worse still, this
value varies with temperature. This has
the effect of producing a zero error con-
sidered at the system input, so that with
short circuit (i.e. zero) input the output
frequency cannot always be set to zero
by R 23 .
This can be balanced out as shown in
figure 3. An extra transistor T 3 is used,
which is biassed by R 19 and R 20 so that
it is permanently in saturation. The de-
gree of saturation is governed by R 20 , and
can be adjusted so that the saturation
voltage equals that which occurs in T 2 .
This voltage is applied to pin 2 of the 74 1 ,
via R 2) , and since the value of R 21 almost
equals the value of ( R j + R 4 ), the voltage
at pin 2 is exactly zero when T 2 is
saturated.
An extra spin-off from this arrangement
is that temperature variations in the satu-
ration voltages of T 2 and T 3 will
approximately track each other, and so
be balanced out.
This modification is shown in the revised
circuit of figure 4, together with several
other changes as follows:
(a) R 2 is reduced to 10 k.
(b) C 2 is increased to lp in order to
improve accuracy for low input
voltage levels, where frequency is
of course low, and a longer time
constant is desirable.
(c) C 4 is increased to 10 n, improving
linearity at high input voltage
levels.
450 — elektor april 1975
a/d converter
S iiiiiSSsiiiliiiii
'• i i
(d) Trimmer R )0 is increased to 22 k,
so that despite the increased
C4 value, ratios of up to 10 KHz/V
can still be set up.
Setting-up procedure
The sequence is as follows. The collectors
ofT 2 and T 3 , and also the voltage input,
are temporarily shorted to earth. The
zero offset pot R 23 is then set to give
zero volts at the 741 output. This adjust-
ment is easier if a 100 k resistor is
temporarily strapped across C, . The
shorts across T 2 and T 3 , and the 100 k
resistor can now be removed.
The pot R 20 is set up for zero output
frequency with zero input voltage. It
should be remembered that since a
negative frequency is meaningless (!) this
setting should be approached by lowering
the input voltage from positive towards
zero, and observing the frequency to
decrease and become zero simultaneous-
ly with input voltage. In some cases it
may be necessary to alter the value of
R19, to compensate for unusual current
gain values encountered in T 3 . In the
same way, R 2 o can be increased to —
say - 470 k.
The next stage is to set the voltage-
frequency conversion factor. The short
Figure 5. Test results.
Figure 6. PCB layout and component pos-
Figure 7. Photograph of setting-up procedure.
Here the DVM has accuracy better than
0.035%, and the frequency counter better
than 0.01%.
circuit on the voltage input (obviously!)
should be removed, and a source of
exactly 1 V connected. To achieve the
best accuracy of which the circuit is
capable, this value should be set up with
a digital voltmeter, or other instrument,
having better than 0.1% accuracy. The
output frequency can then be monitored
on a counter and set up using the Pot R|
to the value desired. The design centre]
value for this circuit is 1 0 KHz/V, but of]
course other values can be set up
required.
Performance Details
Figure 5 gives graphs of the circi
performance after setting up as describ
above. The error in volts, over the whole]
input range 0-1 V is less than 1.5 mV.]
Further, the relative error over the rai
7 mV to 2.5 V is less than 1% of readi
The circuit was also tested without R ;
connected. With R 20 it was set up wil
short circuit input to zero Hz, and the
with Rio to give 10 KHz for 1 V input|
It is to be expected in this case that bol
linearity and temperature stability woul
be worse. Despite this, the accuracy «
± 1% over the range 0.1-1 V was mail
tained.
Best temperature stability will be ol
tained by choosing C 2 , C 3 , and C4 car
fully, poly-carbonate types being 11
commended, (e.g. Siemens MKM). Lii
earity can be further improved by usii
a faster OP-AMP in place of the 709, an
by replacing R4 by a constant-currei
source. However, these sophisticatiol
are only worthwhile if really accural
test gear is available for setting up. |
Z
y
• fl
■
*
1
a
inn
led-displays
elektor april 1975-451
led
displays
In response to numerous requests
from readers we publish this
LED-display chart to enable
constructors to find their way
through the jungle of seven-
segment display data and to
choose alternative displays to
those specified in Elektor
projects.
There are literally dozens of different
seven-segment LED-displays currently
available and it would be prohibitively
expensive to specify and test every suit-
able alternative display for Elektor pro-
jects. This guide is intended to enable the
home constructor to choose such alterna-
tives himself.
This guide is confined to displays with
dual-in-line pin configuration and
common-anode connection as this is the
most popular format and these displays
can be driven by the common 7447 TTL
decoder driver or interfaced with MOS
devices by single-transistor buffer stages
The guide is divided into three sections:
1 . The chart proper.
2. Condensed data on the devices giving
pin connections and important
performance parameters.
3. Hints on choosing devices and calcu-
lating current limiting resistors etc.
The Display chart
This is used in a similar fashion to a
mileage chart in a road atlas. The
required device is first located in the
diagonal list of type numbers (they run
alphabetically by manufacturer, top left
to bottom right). The proposed alterna-
tive is similarly located and if the box
where the corresponding row and column
cross contains a circle than the devices
are direct replacements for each other.
If the box contains a P they are pin com-
patible but the performance data should
be checked to see if a substitution may
be made. For instance, devices of differ-
ent colours may be pin compatible.
Figure 1. The LED-Display chart which may
be used to find pin-compatible and direct
replacement displays.
led-displays
elektor april 1975 — 453
The Data
The data has been extracted from the
relevant manufacturers’ data sheets and
presented in tabular form. An expla-
nation of the symbols used is also given.
Where a particular box in the table con-
tains a dash this means either that the
parameter is not specified or that the
units are not the same as the units used
in the table. For example, some manu-
facturers specify luminous intensity in
millicandelas whereas others specify
brightness in foot-lamberts.
When using the data to choose alterna-
tive devices it is important to check the
physical dimensions of the device. Many
devices have bodies that overhang the
pins, so when using a p.c. board on which
several devices are stacked close together
check that there is sufficient space to
accommodate the width of the device
you propose to use. Some devices have
bodies which are not symmetrical about
the pins but are offset to one side and
these have been excluded from this guide.
Choosing Seven-Segment Displays
When choosing alternative devices to
those specified in a project, note the
following points:
1. Devices that are not directly pin-
compatible may often be used with slight
modification. Many devices have similar
connections for the segment cathodes
and vary only in the connections to
anode or decimal point. The most fre-
quent connection for the common anode
is pin 14. Some devices have additional
anode connections, notably pins 3 and 9.
In some cases the various anode pins are
interconnected inside the package and
are therefore redundant, in other cases
the different anode pins are connected
only to certain segments and must be
connected externally. It is a simple
matter to check which is the case. Con-
nect one of the anode pins to a +5 V
supply via a suitable limiting resistor and
ground each of the segment pins and the
decimal point pin in turn. If all the
segments and the decimal point light then
the anode pins are redundant and only
one of them need be used.
When using a device with redundant
anodes in a circuit board with a single
anode connection simply cut off the
unused anode pins. When using a device
with multiple non-redundant anode con-
nections it is necessary to bend the extra
pins inwards and link them to the pin
used for the common anode connection
to the circuit board. Devices with fewer
anode connections than the device for
which a board was originally intended
present no problems, provided that the
pins which will go into anode connection
holes on the board are NC or may be
cut off.
2. Some devices are available in versions
with left- or right-hand decimal point
and are identical but for the decimal
point connection. In applications where
the decimal point is not required (for
example clocks) the pin may be cut off
if necessary.
3. Having established that a device is, or
Figure 2. Table of pin connections and
performance data for LED-Displays.
Figure 3. General appearance of a seven-
segment LED-Dispiay showing letter desig-
nation of segments and decimal points.
Figure 4. Relationship between light wave-
length and perceived colour.
can be made, compatible with the board,
the next thing to determine is whether
the opto-electronic characteristics of the
device are suitable. One of the most fre-
quent of readers’ queries concerns the
substitution of displays of different
colours to the ones originally specified.
Provided the electrical characteristics are
suitable this is perfectly acceptable but it
must be remembered that yellow and
green devices are often less efficient than
red ones. This is particularly true of
older designs of device which are often
available on the amateur market. Yellow
devices are the least efficient of all. This
effect is fortunately offset to some
extent by the fact that the eye is more
sensitive to yellow and green than it is
to red light, so although yellow and green
displays are often less bright than red
displays operated at the same current the
apparent brightness is not much less.
Nevertheless the difference is often
noticeable.
4. To ensure long device life it is advis-
able to operate displays at or below the
current specified in the If column. As
this may involve recalculation of the
cathode series resistors the formula is
given below
_ Vb ~ Vfe
If
Rk = -
(Vb is the supply voltage; Vf s and If can
be found from the table).
A similar calculation may be performed
for the decimal point cathode resistor by
substituting Vfa for Vf s .
When displays are used in a multiplexed
(strobed) mode then they are only on for
one nth of the time, where n is the num-
ber of displays being multiplexed. Conse-
quently, to maintain the same brightness
as if they were being driven continuously
they will need to be supplied with
n times the current for the time they are
on. Most displays can be strobed at sev-
eral times the continuous forward cur-
rent I av . The formula for calculating the
segment cathode resistors then becomes:
- Vb~ Vfs
nl f
When operating with low supply voltages
(e.g. TTL 5 V) it is advisable to subtract
the saturation voltages of any transistors
used in the multiplex drive circuitry
from the supply voltage when calculating
Rk-
Of course, where it is desired to increase
the current when using an alternative
display it is necessary to ensure that the
circuit can supply the extra current.
454 - elektor april 1975
modulation systems
part 2
modulation
systems
As already announced in Part I,
this instalment deals with Carrier
Position Modulation (CPM), as
well as with frequency and phase modulation. Frequency modulation has
proved to be the best system for VHF broadcasting, while CPM is most
suitable for speech transmission.
Carrier Position Modulation (CPM)
When a speech clipper is used, two
questions arise:
1. What can be gained by using this
system?
2. To what extent is intelligibility
affected?
Experiments with HF clipping on SSB
signals have demonstrated that intelli-
gibility remains good even with infinite
limiting, while the average power is
increased by about 10 dB. If pre-
emphasis is provided ahead of the LF
chain, a further improvement in intelli-
gibility results.
Even with infinite limiting of an SSB
signal there are still some variations in
its amplitude, since the rapid phase
jumps of SSB signals give rise to fre-
quency components outside the trans-
mitted frequency band. Since these
components are filtered out of the
(constant) RF signal, the resultant trans-
mitted signal must contain amplitude
modulation.
If an SSB signal is to be purged of all
amplitude variation, further signal pro-
cessing is needed, and a PLL circuit
happens to be suitable for this. Figure 16
shows the block diagram of an arrange-
ment for producing CPM signals. An
SSB signal is produced from a pre-
emphasised LF input, and after this
signal has been limited it is fed to a PLL
circuit. The VCO in this circuit will
oscillate at the same frequency as the
SSB carrier, but without any amplitude
variations. Component values in the
PLL are chosen to make it unable to
follow rapid phase jumps in the SSB
signal, so that the bandwidth of the
CPM signal is not much greater than
that of the original SSB-signal. Always
provided that care is taken to maintain
intelligibility, remarkably high efficiency
can be achieved with CPM.
Figure 17 shows the relationship be-
tween intelligibility and receiver input
voltage for different modulation sys-
tems. These are based on tests with
sequences of unrelated words, and on
the use of an IF section of the most
suitable form for each system. For the
same degree of intelligibility, the
necessary input voltage with CPM is
less than a third of what is needed with
AM. This means that a CPM signal needs
only one-tenth of the power needed for
a 1 00%-modulated AM-signal, to cover
the same distance. CPM thus offers
higher efficiency for a given transmitter
power.
CPM has only been known for a com-
paratively short time, and amateurs have
experimented very successfully in this
field. Arrangements based on the prin-
ciple outlined in figure 16 are generally
used, but this unfortunately has some
disadvantages. The input signal to the
balanced modulator exhibits amplitude
variations depending, among other
things, on the speaker and the speaker’s
distance from the microphone. The SSB
signal must have greater amplitude than
the carrier injected through Pi, the
purpose of which is to suppress noise
originating from the limiter when there
is no modulation. In producing CPM,
the LF signal must be suitably processed
to avoid these subjective effects, and as
already indicated, this cannot be done
with clipping alone.
The results obtainable with rapid com-
pression are almost the same as those
which can be achieved with logarithmic
amplification, so the latter method is
preferable because of its simplicity.
The block diagram of a CPM transmitter
with LF-signal processing is given in
figure 1 8. A frequency band of 400 Hz
to 3400 Hz from the microphone is
amplified logarithmically and fed to the
balanced modulator. The LF signal now
has only a small degree of amplitude
variation, and it is therefore possible to
inject a higher level of carrier than
would be acceptable without logarithmic
amplification. This results in a better
signal-to-noise ratio for the transmitted
signal.
There is in addition another advantage
to be gained from this configuration,
namely that it can be used for phase
modulation provided the level of carrier
injection is sufficiently high. It can be
shown that the PLL produces a phase-
modulated signal if the balanced-modu-
lator signal emerging from the filter is
smaller than the injected carrier. The
modulation index is a function of the
quotient of these two voltages.
After the VCO signal has been directly
transposed to the desired transmission
frequency, it can be brought up to the
power required by a Class-C amplifier.
CPM can be received in the same way as
SSB, but as an unmodulated carrier
component is available for part of the
time, a PLL can be used. Since a CPM
signal contains no amplitude infor-
mation, amplitude limitation in the
receiver raises no problems, and this
offers a simple means of combating
AM interference in mobile applications.
By way of verification of the advantages
of CPM, experiments were carried out
with an Ultra Low Power transmitter
using the principles of figure 18. A fre-
quency of 27 MHz was used, and the
Table 1.
System
Present application
Efficiency
Future applications
AM
Long-, medium- and shortwave
broadcasting
very low
none
DSSC
SSB
communications networks
high
very high
present applications
CPM
exceptionally high
communications networks
and citizens' radio
FM
NB-FM
PM
high-quality broadcasting
communications networks
usually changed to FM with
integrating networks
high
high
moderate
present applications
continuing
Long-, medium- and short-
wave broadcasting
current applications
continuing
modulation systems
Figure 16. CPM (Carrier Position Modulation)
is effected by processing an infinitely-dipped
SSB signal using a PLL.
Figure 17. Relationship between intelligibility
and receiver input-signal voltage (50% intelli-
gibility is considered inadequate). Note that
the order of merit gives first place to narrow-
band FM (NB - FM), and not SSB.
Figure 18. Intelligibility can be considerably
improved and lining-up simplified by the use
of low-frequency signal processing.
Figure 19. The spectral distribution of an FM
signal can be expressed with Bessel functions.
transmitter power was approx. 20 mW.
In order to compensate, to some extent,
for the unfavourable topographical con-
ditions for VHF propagation - hilly
country - the transmitter and its aerial
were located on a floor of a block of
flats 50 metres up. A loaded aerial rod
was used, and the calculated efficiency
of this combination was 30%, so that
the ERP barely amounted to 6 mW. The
receiver used for this experiment has a
sensitivity of 0.1 /iV with a bandwidth
of 3 kHz and was equipped with a PLL
of the type shown in figure 10 (see
Elektor no. 2).
In spite of the transmitting aerial height,
the optical horizon radius was a bare
7 km. Although reception within this
area was subject to wide fluctuation, it
was observed that the received signal
did not drop below 0.2 #iV.
The limit of receiver sensitivity was
reached at a range of 10km, that is
3 km beyond the optical horizon.
When the transmitter was switched over
to phase modulation, reception at the
optical horizon was observed to have
already become insufficient for reliable
communication.
Frequency Modulation and Phase
Modulation (FM and PM)
When the frequency or the phase of a
carrier is made to vary in accordance
with information, this is known as fre-
456 — elektor april 1975
modulation systems
quency modulation or phase modulation
respectively.
Both satisfy the relationship:
V = V 0 sin(a)hf + m sinco[f t) (4)
in the case of sinusoidal modulation.
The difference between FM and PM lies
in the modulation index m, which is
defined for FM as:
frequency deviation of the RF
carrier with respect to the centre
m _ frequency
modulation frequency v
and with phase modulation m is con-
stant. The expression in (4) can be
expanded to:
V = V 0 J 0 m sin Whf t +
Jim[sin(a>hf+wif)t +
sinfcohf - wjf)t] +
J2m[sin(o;hf + 2coif)t +
sin(cohf - 2 o>if)t] +
J3m[sin(cohf + 3wif)t -
sin(cohf - 3coif)t] + . . .
It can be seen from this that FM and
PM generate a spectrum with infinite
bandwidth. The term J n indicates a
Bessel function of the nth order, whose
magnitude decreases substantially as n
increases. In practice both FM and PM
can therefore be considered to have a
finite bandwidth. Figure 19 shows the
amplitudes of sideband components as
a function of m.
For FM broadcasting, a maximum fre-
21
input signal, RMS (pV) —
quency deviation of 75 kHz and a
maximum modulation frequency of
15 kHz were standardised at the outset.
It follows from (5) that m = 5, and it
can be read off from figure 19 that,
with this modulation index, the relative
amplitude of the seventh-order sideband
is only 0.05. This can be neglected in
most cases because maximum modu-
lation does not occur at 15 kHz in
practice. A rule-of-thumb formula, valid
when m is unity or greater, is:
B w = 2(m + l)fjf max, in which B w
represents the —3 dB bandwidth, m is
the modulation index at the maximum
modulation frequency and fjf max is the
maximum modulation frequency.
The minimum bandwidth needed for
mono FM then works out as:
B^, = 180 kHz. For stereo FM the modu-
lation index has been chosen, on com-
patibility grounds, to enable the mono
bandwidth to be used at the highest
modulation frequency (53 kHz). The
modulation index for the sub-canier
signal conveying the stereo information
can be shown to be 0.6 (as this is less
than unity, the rule-of-thumb formula
does not apply) which results in a 20 dB
deterioration in signal-to-noise ratio. It
can be derived from figure 19 that,
with this low modulation index, the
second-order sideband can be neglected
in practice, as its relative amplitude is
less than 0.05. The bandwidth required
is then no greater than is needed for an
AM system (2.fjf max).
24
Mod. Index =4 j | j
jL
Mod. Index ■ 12 jjyyj
Mi
Mod. Index -24
i
i
i
i
b 4» H
frequency deviation
Figure 20. Relationship between distance
covered and signal-to-noise ratio for AM (1),
FM with 20 kHz deviation (2) and FM w
75 kHz deviation (3).
Figure 22. With low deviations, there is coi
siderable improvement in intelligibility i
low input-signal levels.
Figure 23. These curves clearly show the
superiority of narrow-band FM over AM of
the same bandwidth.
Figure 24. Spectra showing that bandwidth
can be fully utilised by increasing the modu-
lation index for lower frequencies. This is
the case for FM, since the deviation is d<
mined by the amplitude of the modulating
Associated with this lower value of m is
a drop in the maximum signal-to-noise
ratio obtainable, and therefore in the
suppression of both impulsive and
adjacent-channel interference.
One characteristic feature of FM is the
threshold response: this means that the
FM input signal strength must be above
a certain value if it is to be usable. The
threshold value goes down when a lower
modulation index is used. This knowl-
edge is based on research first carried
out in the U.S.A. in the ‘thirties to
determine whether AM of FM would be
best for a reliable police radio network.
Some of the results of these very
extensive researches are reproduced in
figures 20, 21, 22 and 23.
The result of a terrain test is shown
graphically in figure 20. In this case the
transmitter location was fixed, while
the receiver was mobile. Curve 1 is for
AM, Curve 2 for FM with 20 kHz devi-
ation and Curve 3 for FM with 75 kHz
deviation. These curves show that a
higher deviation is needed to give the
high signal-to-noise ratio which hi-fi
demands, but that a price has to be paid
for this in terms of the maximum work-
able range.
Bearing in mind the 20-dB deterioration
in signal-to-noise ratio with the present
stereo system, it is of interest that stereo
broadcasting using two separate trans-
mission links, each with a deviation of
only 20 kHz, would not only give a
better signal-to-noise ratio, but would
also offer a saving in overall bandwidth.
For communication systems, used ex-
clusively for speech transmission, a
maximum modulation frequency of
3 kHz is adequate. Higher values, up to
5 kHz or 6 kHz, are used only when it
is essential to transmit speech of very
high intelligibility.
Figure 2 1 shows a comparison between
a system with 20 kHz deviation and one
with 6 kHz deviation. It will be seen
that the ultimate sensitivity of the
narrow-band system is better by a factor
of 2. In communications networks, a
signal-to-noise ratio of approximately
12 dB is regarded as just usable.
The amateur transmitters’ readability
gradings are also often quoted in intelli-
gibility tests. In figure 22 the amateurs’
intelligibility gradings are plotted
against input signal for different values
of deviation. The narrow-band system is
quite usable with an input of 2 /iV, while
the system with a 20 kHz deviation is un-
readable with this input.
Although limiting sensitivity is consider-
ably better nowadays because of im-
proved reception techniques, this has
no effect on the relationships between
limiting sensitivities with the various
systems.
A comparison between a narrow-band
FM-system and an AM system with the
same bandwidth is given in figure 23.
The curves show the marked superior-
ity of the FM system; this applies not
only to intelligibility, but also to inter-
ference suppression. This points to a
possible alternative for medium waves
which would at least reduce the chaos
prevailing in this band. By re-engineering
the present AM channels for narrow-
band FM with a deviation of 4.5 kHz,
a substantial improvement would be
achieved.
Anyone possessing a short-wave com-
munications receiver equipped for
narrow-band FM reception will find
that, among others, a number of East
European countries are radiating exper-
imental narrow-band FM transmissions,
particularly in the 25-m and 41-m
broadcasting bands. These transmissions
should be of particular service in shed-
ding light on the effect of distortion
caused by selective fading. This distor-
tion seems to be considerably less with
narrow-band FM than with AM and an
envelope detector. As narrow-band FM
is more compatible with AM than is SSB
with a carrier, the change-over to
narrow-band FM could take place
gradually. This move is also advocated
by the fact that, for the same signal-to-
noise ratio, narrow-band FM would
give a 70% saving in transmitter power.
Particularly for narrow-band systems, I
there is a great difference between I
26
"
|— iDi-j
r
' (FM + PM)
-cm-
-1 1-
L-h-J
1
)
£
T
LF
phase- and frequency modulation. Sup-
pose, for example, that the deviation
with FM is 4.5 kHz and the modulation
frequency is 450 Hz, giving a value of
10 for m. This will result in a large
number of sidebands whose main energy
content (J 8 in figure 19) is in the region
of 9 kHz.
With phase modulation m is constant so
that, in the case of narrow-band PM,
as a first approximation two sidebands
will be produced, depending on the
modulation frequency. The sidebands
of AM and narrow-band PM are thus
identical, and this makes it possible to
receive PM with an SSB receiver.
The efficiency of a communication
system is highest when the available
bandwidth is completely filled with
information, and for this reason PM has
a poorer signal-to-noise ratio than FM.
This is illustrated in figure 24, where a
higher modulation index is associated
with lower modulation frequencies.
Figure 25 shows a simple arrangement
for producing a frequency-modulated
oscillation by introducing a varicap into
the LC circuit of a stable oscillator, so
that the oscillator frequency will vary
with modulation. This circuit can be
changed over from FM to PM simply
by feeding the modulation through an
RC section whose cut off frequency is
equal to the highest modulation fre-
quency. As the building of an oscillator
which satisfies Post Office stability
requirements is not exactly a simple
business, crystal-controlled oscillators
are preferable. With these, however,
direct modulation of the oscillator fre-
quency is not possible, as the maximum
deviation cannot be more than 200 parts
per million. However a crystal oscillator
giving phase and frequency modulation
simultaneously may be used, and this
can have the same overall effect for
communications purposes. An example
of such a circuit is given in figure 26.
In many instances, however, the devi-
ation obtained with this circuit will be
too small, and the required deviation
can then be obtained with a frequency-
multiplier circuit. One of many possible
variants of this circuit is shown in
figure 27, and calls for a minimum
number of stages. Crystal-controlled
oscillators XTOi and XT0 2 oscillate at
frequencies f] and f 2 . These frequencies
are multiplied by n and (n + 1 ) respect-
ively and then fed to a mixer stage, the
output of which is tuned to:
fout = ( n+ 1 ) f 2 -nfi.
The two oscillators are modulated with
opposite polarities, via a phase splitter,
giving deviations of Afj and Af 2 respect-
ively, so that the mixer output becomes:
fout + ^h =
(n+ l).(f 2 +Af 2 )-n(f, -Af,).
This can be rearranged to give the
value of the deviation fjj, i.e. :
fh = (n+ 1)-Af 2 +nAfj.
In the case where f j = f 2 and Afi = Af 2 ,
this gives:
f h = (2n+ l)Afi,
with a centre frequency:
fout = •
A practical value for n is three, as this
can be effected with one stage of
multiplication. This gives a seven-fold
multiplication of the deviation, with an
output frequency equal to that of the
crystal oscillators.
When FM signals are detected, imperfect
demodulation causes divergences from
theoretical values (e.g. for interference
suppression), and these divergences
increase as the bandwidth of the system
is reduced. For this reason, special care
should be devoted to the instrumen-
tation of narrow-band FM systems, but
unfortunately the opposite has been
true in the past.
Conclusion
It has been shown that there are only
two modulation systems offering high
efficiency, namely FM and CPM. How-
ever, since CPM conveys no information
on amplitude, this system is only suit-
able for speech transmission. AM is in
every respect the worst system. Although
it appears at first sight to offer economic
advantages, closer study shows up the
disadvantages of AM such as energy
wastage, wavelength clutter and its
contribution to the warming up of the
ionosphere.
FM has rightly been chosen for high-
quality broadcasting, but even FM is
marred by distortion and noise when
new systems, which reduce the modu-
lation index severely, are introduced.
Stereo broadcasting with its information
bandwidth of 53 kHz is a striking
example of this, but it would seem that
yet another step in the wrong direction
is about to be taken with the introduc-
tion of quadraphonic broadcasting. For
a number of quadra systems now being
discussed, a bandwidth of ‘only’ 76 kHz
is needed. In view of the widespread
operation of FM transmitter networks
with a channel spacing of 100 kHz, it
would be preferable to look for tech-
niques which do not call for any increase
in the present bandwidth.
M
Figure 27. With this arrangement, deviation
can be increased without the increase in out-
put frequency which occurs with direct
multiplication.
Modifications to
Additions to
I mprovements on
Corrections in
Circuits published
Elektor
Steam whistle
In the p.c.b. layout for the steam whistle
(Elektor 1, p. 58), the electrolytic ca-
pacitors C 4 , C 7 and C 9 are shown with
the wrong polarity. The negative con-
nections of C 4 and C 7 should be con-
nected to the negative supply line near
the emitter of T 2 ; the positive con-
nection of C 9 should be connected to
the cathode of Di . The circuit diagram
(figure 2) is correct.
feedback
PLL receiver
TUP/TUN tester
quadro in practice
H/L logic probe
jp-tun-dug-dus
elektor april 1975 - 459
TUP
TUP
Tun
Tun
•
UUE
UUE
UUE
Wherever possible in Elektor circuits, transis-
tors and diodes are simply marked 'TUP"
(Transistor, Universal PNP), 'TUN' (Transistor,
Universal NPN), 'DUG' (Diode, Universal Ger-
manium) or ‘DUS' (Diode, Universal Silicon).
This indicates that a large group of similar
devices can be used, provided they meet the
minimum specifications listed above.
For further information, see the article 'TUP-
TUN-DUG-DUS' in Elektor 1, p. 9.
Table 2. Various transistor types that meet the Table 4. Various diodes that meet the DUS or
TUN specifications. DUG specifications.
TUN
BC 107
BC 108
BC 109
BC 147
BC 148
BC 149
BC 171
BC 172
BC 173
BC 182
BC 183
BC 184
BC 207
BC 208
BC 209
BC 237
BC 238
BC 239
BC 317
BC 318
BC 319
BC 347
BC 348
BC 349
BC 382
BC 383
BC 384
BC 407
BC 408
BC 409
BC 413
BC 414
BC 547
BC 548
BC 549
BC 582
BC 583
BC 584
Table 3. Various transistor types that meet the
TUP specifications.
TUP
run
DUE
DUS
DUS
DUG
BA 127
BA 217
BA 218
BA 221
BA 222
BA 317
BA 318
BAX 13
BAY61
1N914
1N4148
OA 85
OA 91
OA 95
AA 1161
Table 5. Minimum specifications for the
BC107, -108, -109 and BC177, -178. -179
families (according to the Pro-Electron
standard). Note that the BC179 does not
necessarily meet the TUP specification
Oc.max = 50 mA).
NPN
PNP
BC 107
BC 177
BC 108
BC 178
BC 109
BC 179
v ce 0
45 V
45 V
20 V
25 V
20 V
20 V
V eb 0
6 V
5 V
5 V
5 V
5 V
5 V
•c
100 m A
100 mA
100 mA
100 mA
100 mA
50 mA
p tot.
300 mW
300 mW
300 mW
300 mW
300 mW
300 mW
f T
150 MHz
130 MHz
mi n
150 MHz
130 MHz
150 MHz
130 MHz
F
10 dB
10 dB
10 dB
10 dB
4 dB
4 dB
The letters after the type number
denote the current gain:
A: a' (P. h fe ) = 125-260
B: a' = 240-500
C: a' = 450-900.
Table 6. Various equivalents for the BC107,
-108, . . . families. The data are those given by
the Pro-Electron standard; individual manu-
facturers will sometimes give better specifi-
cations for their own products.
460 - elektor april 1975
led-level I
;
led-leoel
In general, analogue pointer in-
struments are used for level
indicators. Another method of
indicating amplitudes and power is to use LED's. The advantages of this
system include higher resistance to shock, better legibility from greater
distances and the fact that the response time is unaffected by the mech-
anical time-constant of a conventional meter.
Apart from a practical level meter additional circuits are discussed. The
most important of these is a simple overload indicator.
Figure 1 gives a simple circuit with
which the voltage amplitude on the loud-
speaker output of an amplifier can be
converted into light intensity of lamp L, .
The limiting resistor R, is necessary only
if the lamp can be overdriven by the
amplifier. Of course with a single supply
rail amplifier the circuit of figure 1 must
be connected after the loudspeaker out-
put capacitor. Otherwise the lamp would
be constantly fed from the d.c. mid-point
voltage of the amplifier output stage.
Lamp Lj must bum brightest at maxi-
mum output power. This power is nor-
mally limited by the supply voltage of
the output amplifier. In most cases it can
be said that the maximum output is
obtained if the amplitude of the output
voltage is about 2 volts less than the
supply voltage (also in connection with
increasing distortion). If, for example,
the supply voltage of the amplifier is
24 volts, the maximum swing of the
output voltage will then be about
22 volts peak-to-peak.
The maximum RMS output voltage of
the output stage (from the example) is
half the peak-to-peak voltage divided by
\/2. This is about 7.8 volts. The maxi-
mum voltage of the lamp is 6 volts, so
the surplus of 1.8 volts must drop across
Rj. The resistance value of Ri can now
be calculated by dividing the residual
voltage (1.8 volts) by the 50mA which is
the maximum current for the lamp.
The level indicator
Such a simple system can, at best, give
only an approximate indication of out-
put and its effectiveness depends on
many factors such as ambient lighting
and the eyesight of the individual user.
A much better arrangement is to have
a number of lamps or LEDs which light
in sequence as the voltage is increased.
This is the system used in the LED level
indicator.
The circuit is shown in figure 2. The input
of the circuit is formed by potentio-
meter P, with which the sensitivity is
adjusted. The potentiometer is connected
to the loudspeaker output of the ampli-
fier. If the amplifier is fed asymmetrically
(one supply voltage), potentiometer P!
must be connected after the loudspeakei
output capacitor.
The circuit operates as follows:
I-
«imim mu mu mimcii; uuu r
led-level
Part* list with figure 4
Resistors:
Rl = 1 M
R 2 = 330k
R3.R4= ’Ok
R 5 - 270ft
R 6 = Ik
R7.R8 = 47fi
Pi ■= 1 M, preset potentiometer
Capacitors:
Ci - 0.47/i
C 2 = IOOjU/10 V (see text)
C 3 - 1 00fj/35 V
Semiconductors:
t 1- t 2« t 3 " TUN (above Ufc -
20 V: BC107a)
T 4 = 2N1613
T 5 = 2N2905
D 1 ,D 2 - DUS
Figure 1. The simplest form of level indicator
can be made with a lamp and a resistor. As
the output voltage increases, the lamp will
produce more light. The indication of such a
system is not accurate, and for small voltages
the lamp does not light.
Figure 2. The LED level indicator fitted with
ten LED's. Each time the output voltage in-
creases by about 0.7 V an additional LED will
light up. If the LED's are mounted in line
horizontally or vertically) the result is a ''ther-
mometer" type indication. The length of the
track is an indication of the amplitude of the
output. It is possible to use lamps instead of
LED's. Depending on the type of lamp used,
the load resistors Ri i up to and including R20
may be omitted.
Figure 3. To obtain an indication at low out-
put voltages the anode of diode Di of figure 2
must receive a bias voltage. This is done by
means of an additional adjustment potentio-
meter (P v ), resistor (R v ) and diode (D v ).
Figure 4. If the level indicator must be driven
from a high-output-impedance or low-voltage
source a preamplifier circuit can be used. Its
voltage amplification is 100 or more, depend-
ing on the gain of T3.
Figure 4a. This voltage doubler can replace
diode Di (figure 2) if the indicator fails to
give full deflection. The voltage doubler con-
sists of two diodes (D, and D r ) and two
capacitors (C, and C r ). The doubler can only
be used if the meter has an independent supply.
As appears from the diagram, the loudspeaker
zero and level meter zero (minus terminal of
Ci) are not D.C. connected.
The output voltage of the amplifier
arrives on diode Di via potentiometer Pj .
This diode rectifies the signal positively.
Via D, capacitor C, is charged. If the
voltage across C, increases, there will
come a point where T, conducts. If the
voltage on Ci rises further, transistor T 2
will be driven into conduction via re-
sistor R, .
A resistor and LED are included in the
collector of T 2 . When T 2 conducts,
the LED lights. If the voltage on C, rises
still further, transistor T 3 conducts be-
cause its base is driven via diode D 2 and
resistor R 2 . Now LED D, 2 will also light.
As long as the voltage on capacitor C !
keeps rising, another diode in the chain
D 2 ... D 2 i conducts. Each of the corre-
sponding transistors (T 2 ... T„) and
LEDs (Du ••• D 20 ) also conducts. When
the emitter potential of T, is about
7 volts, all ten LED’s will be lit.
elektor april 1975 — 46 1
If the LED’s are placed in line horizontal-
ly or vertically the result is a light track
whose length is proportional to the out-
put amplitude of the amplifier. Potentio-
meter P 2 in the emitter circuit of Ti
serves to adjust and limit the current.
The indicator responds rapidly to an
increase of the output voltage of the
amplifier. The decay time of the meter
(light track) depends on the value of
capacitor Cj.
At a greater capacitor value the decay
time becomes longer. At the indicated
value for Ci the decay time is about
0.3 seconds.
The circuit may also be fed from higher
voltages. But then the values of R,, up to
and including R 20 must be adapted. The
proper values can be calculated if we
assume that the supply voltage drops at
least 1 .5 volts across a LED and that the
current through the resistors is about
elektor april 1975
40mA. (Ensure that the LED’s used will
stand this current).
If the supply voltage is more than 20 volts
it is not possible to use a TUN. Up to a
supply voltage of 40 volts the TUN’s can
be replaced by BC107a or BC107b.
Instead of LED’s ordinary incandescent
lamps can be used. Their operating
Figure 5. This overload indicator can be used
universally. The input must be connected to
the output of the amplifier before the output
capacitor.
Figure 6. If the level indicator must give an
audiophysiologically corrected indication, this
network may be connected between the loud-
speaker output and the input of the meter.
voltage can best be chosen to equal the
supply voltage. In that case a load
resistor (R n up to and including R 20 ) is
not needed.
A drawback of the circuit of figure 2 is
that the first LED begins to conduct only
after a bias has been built up. If this is
unacceptable, the circuit can be pre-
biased with a resistor, potentiometer, and
diode. Figure 3 gives a detailed drawing
of the input circuit of figure 2 with the
additional components. The bias is ad-
justed with potentiometer P v . Diode D v
serves only to avoid extra loading of the
positive-going loudspeaker signal.
Level preamplifier
If the level indicator must be connected
to a point in the amplifier where there is
not sufficient voltage (and power) to
drive it, the circuit of figure 4 may be
used. This circuit is inserted between the
connecting point in the amplifier and I
potentiometer Pj of figure 2.
The input impedance of the circuit of I
figure 4 is about 270 k. The voltage I
amplification with P] at maximum is I
1 00 X or more. This depends on the gain j
of transistor T 3 . The circuit of figure 4 I
may be connected to supply voltages I
between 1 2 volts and 40 volts. For sup- I
plies higher than 20 volts the TUN’s I
must be replaced by transistors which I
can withstand this voltage (for example
BC107). Furthermore, the operating I
voltage of capacitor C 3 should be at least J
equal to the supply voltage.
If the supply voltage for the circuit of I
figure 4 is less than 20 volts, the level I
meter cannot be fully driven under nor- I
mal conditions. To achieve this, diode D! |
(from figure 2) must be replaced by a I
voltage doubler, so that capacitor C] (of I
figure 2) receives about twice the voltage I
(see figure 4a).
Overload indicator
It can be quite handy if a power amplifier I
is provided with a device that indicates I
when the amplifier is overdriven: an I
overload indicator. Figure 5 gives a prac- |
tical example. The input is connected to I
the output of the amplifier. Since we are
now concerned with overdrive, the input I
must be connected before the loud- I
speaker elco.
The threshold level of the overload I
indicator may be adjusted by potentio- I
meter P! . This adjustment must be such I
that if a certain level is exceeded, the I
fiA 741 switches, and produces a positive I
voltage. This voltage drives transistor T i .
The emitter circuit of Ti includes an I
incandescent lamp or LED which then I 1
lights. I *
The overload indicator of figure 5 can I
also be used for higher voltages (up to I
37 volts). The value of resistor R 3 must I
be increased in proportion with the 1
higher supply voltage. To ensure the I
survival of the IC, the input voltage I
should not be more than the supply I
voltage. For this reason an extra resistor I
(Rx) of 10 ... 22k in the input lead may I
be needed.
Physiological correction
If the level indicator must give an audio- I
physiologically corrected indication, the I
network of figure 6 can be connected I
oetween the meter and the loudspeaker I
output. This network gives an attenua- I
tion of about 4 X. If the input voltage is I
then insufficient to drive the meter to I
maximum indication, there are two pos- I
sible solutions. The voltage doubler of I
figure 4a can be used, or alternatively I _
the circuit of figure 4 can be connected I s
between the correction network output ■ Li
and the input of the level indicator. In I*
that case potentiometer P! and the I ■
capacitors Ci and C 2 can be omitted- “
from the circuit of figure 4.
With the audio-physiologically corrected
level meter it is necessary to use an over-
load indicator, because it is impossible
to see when the amplifier is giving its
peak power. M
imiifMS'imPisfi;
elektor april 1975 - 463
mHRKBT
Quadrophonic cartridge
from Elac
A new range of pickup cartridges
manufactured by Electro-acoustic
GmbH of West Germany is now
available in Great Britain.
Illustrated is the
ELAC STS 655- D4 cartridge,
which is designed for playing
quadrophonic carrier discs. It is
fitted with a parabolically ground
Shibata diamond stylus and will
track at up to 50 kHz. The
cartridge may also be used with
normal stereo or matrix quadro-
phonic discs. Elac cartridges range
in price from £10- £49.
Camouflaged Speakers
For those who wish their Hi-Fi to
be unobtrusive, the ECHONICA
speakers from Japan may be the
answer. Having the appearance of
a picture in a frame only
1 Vi inches deep, these speakers are
designed for wall mounting. The
'canvas’ of the picture is the
loudspeaker diaphragm and a
range of 60 pictures is available.
The price is £47.00 a pair plus
V.A.T.
‘I
Low-cost 50-ohm
Sweep/Function
Generator
A new sweep/function generator
is available from Dana Electronics
Ltd. The model 196 A offers sine,
triangle, square, pulse, ramp and
sweep waveforms over the range
0.1 Hz-1 MHz, in seven ranges.
The generator will provide 10 V
open-circuit or 20 V into a
50 ohm load. Attenuation up to
70 dB is provided in two 20 dB
fixed steps and a 30 dB variable.
An internal sweep generator will
sweep the output frequency over
up to three decades, with sweep
rates from 1 mS to 10 S. A
separate TTL compatible square-
wave output is provided. Size is
187 x73 x 216 mm (7.5 x 2.9 x
8.6 inches) and it weighs less than
1 kg. Price is £195.
Logic Probe
A new TTL/DTL logic probe is
available from Intercontinental
Components Ltd. Readout is by
four LED’s. H and L to indicate
high or low logic states at the
input and Q and Q, to indicate the
state of a storage latch, which
toggles on a positive transition at
the input The probe derives its
45 mA supply current from the
circuit under test and is reverse
polarity protected. Probe input
current is 2.4 mA max. and
response time is 50 ns. The
one-off price is £1 1.50.
New family of low-power
TTL devices
National Semiconductor have
announced the start of volume
production of a new range of
low-power TTL devices known as
54 LS/74 LS Low-Power
Schottky or LPS. The first nine
types, 74 LS00, 01, 03, 04, 10,
12, 20, 22, and 30 are now
available in quantity. Suggested
resale unit prices in lots of 100’s
are £0.20 for all gates, except
the 74LS04 which is priced at
£0.22. It is anticipated that, by
mid-1976, all of the popular
circuits that are now in the
standard 54/74 family will have
been duplicated.
The 54LS/74LS devices are
claimed to have the best speed-to-
power ratio of any high-speed
logic family on the market
Compared with standard
TTL devices, low-power
Schottky logic dissipates only
one-fifth the amount of power
(2 mW per gate) while making no
sacrifice in operating speed.
Low-power Schottky will replace
most high-speed TTL logic, and
can be used in some Schottky
TTL applications, as well as
standard TTL, since the LS series
has dynamic characteristics that
closely approximate those of the
standard 54/74 TTL It is possible
to remove a 7400 device and
insert a 74 LS 00 device in its
place and obtain the same speed
with lower power consumption.
Compact digital
multimeter
The ‘Danameter’ is an almost
pocket-sized digital multimeter
from Dana Electronics Ltd. The
instrument is powered by a single
9 V transistor radio battery,
which should last for up to a year
of normal use, and has a 3!4 digit
liquid crystal display that adjusts
itself to ambient light levels.
Sixteen ranges are selectable by
means of a single, 18-position
switch, with two positions for
‘off’ and battery test. The case is
moulded in high-impact a-b.s.
plastic and the manufacturers
claim that the meter will survive
bench-high drops and drastic
electrical overloads such as 250 V
on the ohms ranges. Ranges are 2,
20, 200 and 1000 D.C. and A.C.
20 p A, 2 mA, 200 mA and 2A
D.C. 200 ohms, 20 k and 2 M.
Dimension are 102 x 184 x
57 mm. (4 x 7.25 x 2.25 inches)
and the weight is 0.45 kg (1 lb.).
Price of the basic Danameter is
£99.50.
Versatile Multimeter
A new multimeter is available
from Metrawatt U.K. Ltd. The
Unigor A42 Multimeter has a total
of 30 measuring ranges for
A.C./D.G current and voltage,
and resistance.
Ranges are: D.C volts, 60 mV-
12 kV, accuracy
± 1% F.S.D.
A.C volts, 6 V-
12 kV, accuracy
± 1% F.S.D.
/' N
D.C amps, 60 flA-
30 A, accuracy
± 1% F.S.D.
A.C. amps, 0.6 mA-
30 A, accuracy
± 1.5% F.S.D.
ohms, 0-1 M, accu-
racy ± 1.5% full
± 6% true value at
mid-scale.
The instrument is shock-proof
and overload protected,
measures 212 x 110 x 82 mm
(8.5 x 4.4 x 3.3 inches) and
weighs 1 kg (2.2 lb.). A range
of accessories is available. One-off
price is £55.
New Varactor Diodes
The ITT Components Group is
introducing a number of new
types of varactor diode. This VUE
series consists of an improved
range of step-recovery mesa
diodes with a screened epitaxial
structure. These components can
be used in frequency multipliers
with output frequencies from
4-8 GHz to 10-14 GHz.
The VUE series comprises the
tuning diodes VSA413H,
VSA417H, VYA413H and
VYA417H, all of which are
suitable for the VHF and UHF
bands as well as for microwave
frequencies. The minimum quality
factor at 50 MHz is 1000 for the
41 3H types and 800 for the
417H types.
r
ELECTROVALOE LTdI
Telephone (061 1 432 4946. Shop hours: 0*ly 9-6.30 p.m.. 9-1 Sots
U.S-A. CUSTOMERS »re inuitsd 10 conwct ELECTROVALUE AMERICA.
Hannover
Messe’Tr l S J
I6.-Z4. April # l#
Wir stellen aus:
Elektor
MOS Clock
and
iponc
in stock
We specialise in high-quality components at
unbeatable prices.
Full range of Elektor boards ex-stock.
MOS Clock (two boards) £ 1-90
TV Sound £1-30
High quality Disc Preamp £ 0-85
Aerial Amplifier £ 0-85
ELECTRONICS
283 Edgware Road, London W2.
Tel. 01-262 8614
ELEKTOR VERLAG GMBH
D-5133 Gangelt 1, W-Germany Tel. (W. Germany) 02454-5055
Hours of business 9.30 - 6.00
Monday to Saturday
Now you can
change record-speeds
without changing
record-speeds.
WeVe done away with the turntable and pick-up arm.
old turntable speed-control, on The tracking error of the practically frictionless
this very advanced Philips GA209 pick-up arm is very small,
record deck. Side thrust compensation is adjustable for all
Simply by placing a record on playing weights for both spherical and elliptical styli.
the turntable the correct speed is The top cartridge from the Super M range, the
electronically chosen and the GP412, is supplied as standard,
pick-up lowered gently into the
run-in groove.
At the end of the record the turntable stops and
the arm returns to the rest.
This facility ensures that both the record and
stylus are fully protected.
In manual operation, the pick-up can be
positioned over the grooves and lowered by means
of a touch control.
The mechanism permits very accurate positioning.
Controlled by a servo motor via electronic touch
controls, it can be operated whether the deck is used
manually or as a fully automatic deck.
Electronic control makes sure that the turntable
speed is kept constant.
Separate fine speed controls for 33 '/j and 45 rpm.
allow the record to be tuned to the pitch of any
musical instrument.
The photo-electric stop switch is completely
soundless and frictionless.
High stability and insulation against shocks and
vibration are ensured by the floating suspension of the
PHILIPS
Sim ply years ahead
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